| 1 | .th |
| 2 | .(l C |
| 3 | .b "Spice VAX Version 2X.x User's Guide" |
| 4 | .sp 0.2i |
| 5 | R.Dowell, A.R.Newton, D.O.Pederson |
| 6 | Department of Electrical Engineering and Computer Sciences |
| 7 | University of California |
| 8 | Berkeley, Ca., 94720 |
| 9 | .sp 0.2i |
| 10 | .)l |
| 11 | .pp |
| 12 | Spice is a general-purpose circuit simulation program for nonlinear dc, |
| 13 | nonlinear transient, and linear ac analyses. Circuits may contain resistors, |
| 14 | capacitors, inductors, mutual inductors, independent voltage and current |
| 15 | sources, four types of dependent sources, transmission lines, and the four most |
| 16 | common semiconductor devices: diodes, bjts, jfets, and mosfets. |
| 17 | .pp |
| 18 | Spice has built-in models for the semiconductor devices, and the user need |
| 19 | specify only the pertinent model parameter values. The model for the bjt is |
| 20 | based on the integral charge model of Gummel and Poon; however, if the Gummel- |
| 21 | Poon parameters are not specified, the model reduces to the simpler Ebers-Moll |
| 22 | model. In either case, charge storage effects, ohmic resistances, and a |
| 23 | current-dependent output conductance may be included. The diode model can be |
| 24 | used for either junction diodes or schottky barrier diodes. The jfet model is |
| 25 | based on the fet model of Shichman and Hodges. The model for the mosfet is |
| 26 | based on the Frohman-Grove model; however, channel-length modulation, |
| 27 | subthreshold conduction, and some short-channel effects are included. |
| 28 | .pp |
| 29 | Note that the mosfet model parameter lambda has been changed to express |
| 30 | channel length modulation in meters/volt in this version of spice. |
| 31 | .bp |
| 32 | .sh 1 "TYPES OF ANALYSIS" |
| 33 | .sp 0.2i |
| 34 | .sh 2 "dc analysis" |
| 35 | .pp |
| 36 | The dc analysis portion of spice determines the dc operating point of the |
| 37 | circuit with inductors shorted and capacitors opened. A dc analysis is |
| 38 | automatically performed prior to a transient analysis to determine the transient |
| 39 | initial conditions, and prior to an ac small-signal analysis to determine the |
| 40 | linearized, small-signal models for nonlinear devices. If requested, the dc |
| 41 | small-signal value of a transfer function (ratio of output variable to input |
| 42 | source), input resistance, and output resistance will also be computed as a |
| 43 | part of the dc solution. The dc analysis can also be used to generate dc |
| 44 | transfer curves: a specified independent voltage or current source is stepped |
| 45 | over a user-specified range and the dc output variables are stored for each |
| 46 | sequential source value. If requested, spice also will determine the dc |
| 47 | small-signal sensitivities of specified output variables with respect to circuit |
| 48 | parameters. The dc analysis options are specified on the .dc, .tf, .op, |
| 49 | and .sens control cards. |
| 50 | .pp |
| 51 | If one desires to see the small-signal models for nonlinear devices |
| 52 | in conjunction with a transient analysis operating point, then the '.op' |
| 53 | card must be provided. The dc bias conditions will be identical for each |
| 54 | case, but the more comprehensive operating point information is not available |
| 55 | to be printed when transient initial conditions are computed. |
| 56 | .sp 0.2i |
| 57 | .sh 2 "ac small-signal analysis" |
| 58 | .pp |
| 59 | The ac small-signal portion of spice computes the ac output variables as a |
| 60 | function of frequency. The program first computes the dc operating point of |
| 61 | the circuit and determines linearized, small-signal models for all of the |
| 62 | nonlinear devices in the circuit. The resultant linear circuit is then analyzed |
| 63 | over a user-specified range of frequencies. The desired output of an ac small- |
| 64 | signal analysis is usually a transfer function (voltage gain, transimpedance, |
| 65 | etc). If the circuit has only one ac input, it is convenient to set that input |
| 66 | to unity and zero phase, so that output variables have the same value as the |
| 67 | transfer function of the output variable with respect to the input. |
| 68 | .pp |
| 69 | The generation of white noise by resistors and semiconductor devices can |
| 70 | also be simulated with the ac small-signal portion of spice. Equivalent noise |
| 71 | source values are determined automatically from the small-signal operating |
| 72 | point of the circuit, and the contribution of each noise source is added at a |
| 73 | given summing point. The total output noise level and the equivalent input |
| 74 | noise level are determined at each frequency point. The output and input noise |
| 75 | levels are normalized with respect to the square root of the noise bandwidth |
| 76 | and have the units volts/rt hz or amps/rt hz. The output noise and equivalent |
| 77 | input noise can be printed or plotted in the same fashion as other output |
| 78 | variables. No additional input data is necessary for this analysis. |
| 79 | .pp |
| 80 | Flicker noise sources can be simulated in the noise analysis by including |
| 81 | values for the parameters kf and af on the appropriate device model cards. |
| 82 | .pp |
| 83 | The distortion characteristics of a circuit in the small-signal mode can |
| 84 | be simulated as a part of the ac small-signal analysis. The analysis is |
| 85 | performed assuming that one or two signal frequencies are imposed at the input. |
| 86 | .pp |
| 87 | The frequency range and the noise and distortion analysis parameters are |
| 88 | specified on the .ac, .noise, and .distortion control lines. |
| 89 | .sp 0.2i |
| 90 | .sh 2 "transient analysis" |
| 91 | .pp |
| 92 | The transient analysis portion of spice computes the transient output |
| 93 | variables as a function of time over a user-specified time interval. The |
| 94 | initial conditions are automatically determined by a dc analysis. All sources |
| 95 | which are not time dependent (for example, power supplies) are set to their dc |
| 96 | value. For large-signal sinusoidal simulations, a fourier analysis of the |
| 97 | output waveform can be specified to obtain the frequency domain fourier |
| 98 | coefficients. The transient time interval and the fourier analysis options are |
| 99 | specified on the .tran and .fourier control lines. |
| 100 | .sp 0.2i |
| 101 | .sh 2 "analysis at different temperatures" |
| 102 | .pp |
| 103 | All input data for spice is assumed to have been measured at 25 deg c |
| 104 | (298 deg k). The simulation also assumes a nominal temperature of 25 deg c. |
| 105 | The circuit can be simulated at other temperatures by using a .temp control |
| 106 | line. |
| 107 | .pp |
| 108 | Temperature appears explicitly in the exponential terms of the bjt and |
| 109 | diode model equations. In addition, saturation currents have a built-in |
| 110 | temperature dependence. The temperature dependence of the saturation current |
| 111 | in the bjt models is determined by: |
| 112 | .(l |
| 113 | js(T1) = js(T0)*((T1/T0)**pt)*exp(q*Eg*(T1-T0)/(k*T1*T0)) |
| 114 | .)l |
| 115 | where k is boltzmans constant, q is the electronic charge, Eg is the energy |
| 116 | gap which is a model parameter, and pt is the saturation current |
| 117 | temperature exponent (also a model parameter, and usually equal to 3). The |
| 118 | temperature dependence of forward and reverse beta is according to the formula: |
| 119 | .(l |
| 120 | beta(T1)=beta(T0)*(T1/T0)**tb |
| 121 | .)l |
| 122 | where T1 and T0 are in degrees kelvin, and tb is a user-supplied model |
| 123 | parameter. Temperature effects on beta are carried out by appropriate |
| 124 | adjustment to the values of bf, jle, br, and jlc. Temperature dependence of the |
| 125 | saturation current in the junction diode model is determined by: |
| 126 | .(l |
| 127 | is(T1) = is(T0)*((T1/T0)**(pt/n))*exp(q*Eg*(T1-T0)/(k*n*T1*T0)) |
| 128 | .)l |
| 129 | where n is the emission coefficient, which is a model parameter, and the other |
| 130 | symbols have the same meaning as above. Note that for schottky barrier diodes, |
| 131 | the value of the saturation current temperature exponent, pt, is usually 2. |
| 132 | .pp |
| 133 | Temperature appears explicitly in the value of junction potential, phi, |
| 134 | for all the device models. The temperature dependence is determined by: |
| 135 | .(l |
| 136 | phi(temp) = k*temp/q*log(Na*Nd/Ni(temp)**2) |
| 137 | .)l |
| 138 | where k is boltzmans constant, q is the electronic charge, Na is the acceptor |
| 139 | impurity density, Nd is the donor impurity density, Ni is the intrinsic |
| 140 | concentration, and Eg is the energy gap. |
| 141 | .pp |
| 142 | Temperature appears explicitly in the value of surface mobility, uo, for |
| 143 | the mosfet model. The temperature dependence is determined by: |
| 144 | .(l |
| 145 | uo(temp) = uo(tnom)/(temp/tnom)**(1.5) |
| 146 | .)l |
| 147 | .pp |
| 148 | The effects of temperature on resistors is modeled by the formula: |
| 149 | .(l |
| 150 | value(temp) = value(tnom)*(1+tc1*(temp-tnom)+tc2*(temp-tnom)**2)) |
| 151 | .)l |
| 152 | where temp is the circuit temperature, tnom is the nominal temperature, and |
| 153 | tc1 and tc2 are the first- and second-order temperature coefficients. |
| 154 | .sp 0.5i |
| 155 | .sh 1 "CONVERGENCE" |
| 156 | .sp 0.2i |
| 157 | .pp |
| 158 | Both dc and transient solutions are obtained by an iterative process which |
| 159 | is terminated when both of the following conditions hold: |
| 160 | .sp 0.2i |
| 161 | .ip 1) |
| 162 | The nonlinear branch currents converge to within a tolerance of |
| 163 | 0.1 percent or 1 picoamp (1.0e-12 amp), whichever is larger. |
| 164 | .ip 2) |
| 165 | The node voltages converge to within a tolerance of 0.1 percent |
| 166 | or 1 microvolt (1.0e-6 volt), whichever is larger. |
| 167 | .pp |
| 168 | Although the algorithm used in spice has been found to be very reliable, in |
| 169 | some cases it will fail to converge to a solution. When this failure occurs, |
| 170 | the program will print the node voltages at the last iteration and terminate |
| 171 | the job. In such cases, the node voltages that are printed are not necessarily |
| 172 | correct or even close to the correct solution. |
| 173 | .pp |
| 174 | Failure to converge in the dc analysis is usually due to an error in |
| 175 | specifying circuit connections, element values, or model parameter values. |
| 176 | Regenerative switching circuits or circuits with positive feedback probably |
| 177 | will not converge in the dc analysis unless the 'off' option is used for some |
| 178 | of the devices in the feedback path, or the .nodeset card is used to force the |
| 179 | circuit to converge to the desired state. |
| 180 | .sp 0.2i |
| 181 | .bp |
| 182 | .sh 1 "INPUT FORMAT" |
| 183 | .sp 0.2i |
| 184 | .pp |
| 185 | The input format for spice is of the free format type. Fields on a card |
| 186 | are separated by one or more blanks, a comma, an equal (=) sign, or a left or |
| 187 | right parenthesis; extra spaces are ignored. A card may be continued by |
| 188 | entering a + (plus) in column 1 of the following card; spice continues reading |
| 189 | beginning with column 2. |
| 190 | .pp |
| 191 | A name field must begin with a letter (a through z) and cannot contain |
| 192 | any delimiters. Only the first eight characters of the name are used. |
| 193 | .pp |
| 194 | A number field may be an integer field (12, -44), a floating point field |
| 195 | (3.14159), either an integer or floating point number followed by an integer |
| 196 | exponent (1e-14, 2.65e3), or either an integer or a floating point number |
| 197 | followed by one of the following scale factors: |
| 198 | .sp 0.2i |
| 199 | .TS |
| 200 | center; |
| 201 | l l l l l. |
| 202 | t=1e12 g=1e9 meg=1e6 k=1e3 mil=25.4e-6 |
| 203 | m=1e-3 u=1e-6 n=1e-9 p=1e-12 f=1e-15 |
| 204 | .TE |
| 205 | .sp 0.2i |
| 206 | Letters immediately following a number that are not scale factors are ignored, |
| 207 | and letters immediately following a scale factor are ignored. Hence, 10, 10v, |
| 208 | 10volts, and 10hz all represent the same number, and m, ma, msec, and mmhos all |
| 209 | represent the same scale factor. Note that 1000, 1000.0, 1000hz, 1e3, 1.0e3, |
| 210 | 1khz, and 1k all represent the same number. |
| 211 | .bp |
| 212 | .sh 1 "CIRCUIT DESCRIPTION" |
| 213 | .pp |
| 214 | The circuit to be analyzed is described to spice by a set of element |
| 215 | cards, which define the circuit topology and element values, and a set of |
| 216 | control cards, which define the model parameters and the run controls. The |
| 217 | first card in the input deck must be a title card, and the last card must be |
| 218 | a .end card. The order of the remaining cards is arbitrary (except, of course, |
| 219 | that continuation cards must immediately follow the card being continued). |
| 220 | .pp |
| 221 | Each element in the circuit is specified by an element card that contains |
| 222 | the element name, the circuit nodes to which the element is connected, and the |
| 223 | values of the parameters that determine the electrical characteristics of the |
| 224 | element. The first letter of the element name specifies the element type. |
| 225 | The format for the spice element types is given in what follows. The strings |
| 226 | 'xxxxxxx', 'yyyyyyy', and 'zzzzzzz' denote arbitrary alphanumeric strings. For |
| 227 | example, a resistor name must begin with the letter r and can contain from one |
| 228 | to eight characters. Hence, r, r1, rse, rout, and r3ac2zy are valid resistor |
| 229 | names. |
| 230 | .pp |
| 231 | Data fields that are enclosed in lt and gt signs '< >' are optional. All |
| 232 | indicated punctuation (parentheses, equal signs, etc.) are required. With |
| 233 | respect to branch voltages and currents, spice uniformly uses the associated |
| 234 | reference convention (current flows in the direction of voltage drop). |
| 235 | .pp |
| 236 | Nodes must be nonnegative integers but need not be numbered sequentially. |
| 237 | The datum (ground) node must be numbered zero. The circuit cannot contain a |
| 238 | loop of voltage sources and/or inductors and cannot contain a cutset of current |
| 239 | sources and/or capacitors. Each node in the circuit must have a dc path to |
| 240 | ground. Every node must have at least two connections except for transmission |
| 241 | line nodes (to permit unterminated transmission lines) and mosfet substrate |
| 242 | nodes (which have two internal connections anyway). |
| 243 | .sh 1 "TITLE CARD, COMMENT CARDS AND .END CARD" |
| 244 | .sp 0.2i |
| 245 | .sh 2 "title card" |
| 246 | .sp 0.2i |
| 247 | .b "Examples:" |
| 248 | .(l |
| 249 | power amplifier circuit |
| 250 | test of CAM cell |
| 251 | .)l |
| 252 | .pp |
| 253 | This card must be the first card in the input deck. Its contents are |
| 254 | printed verbatim as the heading for each section of output. |
| 255 | .sh 2 ".end card" |
| 256 | .sp 0.2i |
| 257 | .b "Examples:" |
| 258 | .(l |
| 259 | .end |
| 260 | .)l |
| 261 | .pp |
| 262 | This card must always be the last card in the input deck. Note that the |
| 263 | period is an integral part of the name. |
| 264 | .sp 0.2i |
| 265 | .sh 2 "comment card" |
| 266 | .sp 0.2i |
| 267 | .b "General form:" |
| 268 | .(l |
| 269 | * <any comment> |
| 270 | .)l |
| 271 | .b "Examples:" |
| 272 | .(l |
| 273 | * rf=1k gain should be 100 |
| 274 | * May the Force be with my circuit |
| 275 | .)l |
| 276 | .pp |
| 277 | The asterisk in the first column indicates that this card is a |
| 278 | comment card. Comment cards may be placed anywhere in the circuit description. |
| 279 | .bp |
| 280 | .sh 1 "ELEMENT CARDS" |
| 281 | .sp 0.2i |
| 282 | .sh 2 "resistors" |
| 283 | .sp 0.2i |
| 284 | .b "General form:" |
| 285 | .(l |
| 286 | rxxxxxxx n1 n2 value <tc=tc1<,tc2>> |
| 287 | .)l |
| 288 | .b "Examples:" |
| 289 | .(l |
| 290 | r1 1 2 100 |
| 291 | rc1 12 17 1k tc=0.001,0.015 |
| 292 | .)l |
| 293 | .pp |
| 294 | N1 and n2 are the two element nodes. Value is the resistance (in ohms) |
| 295 | and may be positive or negative but not zero. Tc1 and tc2 are the (optional) |
| 296 | temperature coefficients; if not specified, zero is assumed for both. The |
| 297 | value of the resistor as a function of temperature is given by: |
| 298 | .(l |
| 299 | value(temp) = value(tnom)*(1+tc1*(temp-tnom)+tc2*(temp-tnom)**2)) |
| 300 | .)l |
| 301 | .sp 0.4i |
| 302 | .sh 2 "capacitors and inductors" |
| 303 | .sp 0.2i |
| 304 | .b "General form:" |
| 305 | .(l |
| 306 | cxxxxxxx n+ n- value <ic=incond> |
| 307 | lyyyyyyy n+ n- value <ic=incond> |
| 308 | .)l |
| 309 | .sp 0.2i |
| 310 | .b "Examples:" |
| 311 | .(l |
| 312 | cbyp 13 0 1uf |
| 313 | cosc 17 23 10u ic=3v |
| 314 | llink 42 69 1uh |
| 315 | lshunt 23 51 10u ic=15.7ma |
| 316 | .)l |
| 317 | .pp |
| 318 | N+ and n- are the positive and negative element nodes, respectively. |
| 319 | Value is the capacitance in farads or the inductance in henries. |
| 320 | .pp |
| 321 | For the capacitor, the (optional) initial condition is the initial |
| 322 | time-zero) value of capacitor voltage (in volts). For the inductor, the (option |
| 323 | initial condition is the initial (time-zero) value of inductor current (in |
| 324 | amps) that flows from n+, through the inductor, to n-. Note that the initial |
| 325 | conditions (if any) apply 'only' if the uic option is specified on the .tran |
| 326 | card. |
| 327 | .sh 2 "coupled (mutual) inductors" |
| 328 | .sp 0.2i |
| 329 | .b "General form:" |
| 330 | .(l |
| 331 | kxxxxxxx lyyyyyyy lzzzzzzz value |
| 332 | .)l |
| 333 | .b "Examples:" |
| 334 | .(l |
| 335 | k43 laa lbb 0.999 |
| 336 | kxfrmr l1 l2 0.87 |
| 337 | .)l |
| 338 | .pp |
| 339 | lyyyyyyy and lzzzzzzz are the names of the two coupled inductors, and |
| 340 | value is the coefficient of coupling, k, which must be greater than 0 and less |
| 341 | than or equal to 1. Using the 'dot' convention, place a 'dot' on the first |
| 342 | node of each inductor. |
| 343 | .sp 0.2i |
| 344 | .sh 2 "transmission lines (lossless)" |
| 345 | .sp 0.2i |
| 346 | .b "General form:" |
| 347 | .(l |
| 348 | txxxxxxx n1 n2 n3 n4 z0=value <td=value> <f=freq <nl=nrmlen>> |
| 349 | + <ic=v1,i1,v2,i2> |
| 350 | .)l |
| 351 | .sp 0.2i |
| 352 | .b "Examples:" |
| 353 | .(l |
| 354 | t1 1 0 2 0 z0=50 td=10ns |
| 355 | .)l |
| 356 | .pp |
| 357 | N1 and n2 are the nodes at port 1; n3 and n4 are the nodes at port 2. |
| 358 | Z0 is the characteristic impedance. The length of the line may be expressed in |
| 359 | either of two forms. The transmission delay, td, may be specified directly (as |
| 360 | td=10ns, for example). Alternatively, a frequency f may be given, together |
| 361 | with nl, the normalized electrical length of the transmission line with respect |
| 362 | to the wavelength in the line at the frequency f. If a frequency is specified |
| 363 | but nl is omitted, 0.25 is assumed (that is, the frequency is assumed to be the |
| 364 | quarter-wave frequency). Note that although both forms for expressing the line |
| 365 | length are indicated as optional, one of the two must be specified. |
| 366 | .pp |
| 367 | Note that this element models only one propagating mode. If all four |
| 368 | nodes are distinct in the actual circuit, then two modes may be excited. To |
| 369 | simulate such a situation, two transmission-line elements are required. (see |
| 370 | the example in Appendix A for further clarification.) |
| 371 | .pp |
| 372 | The (optional) initial condition specification consists of the voltage |
| 373 | and current at each of the transmission line ports. Note that the initial |
| 374 | conditions (if any) apply 'only' if the uic option is specified on the .tran |
| 375 | card. |
| 376 | .pp |
| 377 | One should be aware that spice will use a transient time-step which |
| 378 | does not exceed 1/2 the minimum transmission line delay. Therefore very |
| 379 | short transmission lines (compared with the analysis time frame) will cause |
| 380 | long run times. |
| 381 | .sh 2 "linear dependent sources" |
| 382 | .pp |
| 383 | Spice allows circuits to contain linear dependent sources characterized by |
| 384 | any of the four equations |
| 385 | .sp 0.2i |
| 386 | i=g*v v=e*v i=f*i v=h*i |
| 387 | .sp 0.2i |
| 388 | where g, e, f, and h are constants representing transconductance, voltage gain, |
| 389 | current gain, and transresistance, respectively. Note: a more complete |
| 390 | description of dependent sources as implemented in spice is given in Appendix B. |
| 391 | .sp 0.2i |
| 392 | .sh 2 "linear voltage-controlled current sources" |
| 393 | .sp 0.2i |
| 394 | .b "General form:" |
| 395 | .(l |
| 396 | gxxxxxxx n+ n- nc+ nc- value |
| 397 | .)l |
| 398 | .sp 0.2i |
| 399 | .b "Examples:" |
| 400 | .(l |
| 401 | g1 2 0 5 0 0.1mmho |
| 402 | .)l |
| 403 | .pp |
| 404 | N+ and n- are the positive and negative nodes, respectively. Current flow |
| 405 | is from the positive node, through the source, to the negative node. Nc+ and |
| 406 | nc- are the positive and negative controlling nodes, respectively. Value is |
| 407 | the transconductance (in mhos). |
| 408 | .sp 0.2i |
| 409 | .sh 2 "linear voltage-controlled voltage sources" |
| 410 | .sp 0.2i |
| 411 | .b "General form:" |
| 412 | .(l |
| 413 | exxxxxxx n+ n- nc+ nc- value |
| 414 | .)l |
| 415 | .sp 0.2i |
| 416 | .b "Examples:" |
| 417 | .(l |
| 418 | e1 2 3 14 1 2.0 |
| 419 | .)l |
| 420 | .pp |
| 421 | N+ is the positive node, and n- is the negative node. Nc+ and nc- are the |
| 422 | positive and negative controlling nodes, respectively. Value is the voltage |
| 423 | gain. |
| 424 | .sp 0.2i |
| 425 | .sh 2 "linear current-controlled current sources" |
| 426 | .sp 0.2i |
| 427 | .b "General form:" |
| 428 | .(l |
| 429 | fxxxxxxx n+ n- vnam value |
| 430 | .)l |
| 431 | .sp 0.2i |
| 432 | .b "Examples:" |
| 433 | .(l |
| 434 | f1 13 5 vsens 5 |
| 435 | .)l |
| 436 | .pp |
| 437 | N+ and n- are the positive and negative nodes, respectively. Current flow |
| 438 | is from the positive node, through the source, to the negative node. Vnam is |
| 439 | the name of a voltage source through which the controlling current flows. The |
| 440 | direction of positive controlling current flow is from the positive node, |
| 441 | through the source, to the negative node of vnam. Value is the current gain. |
| 442 | .sp 0.2i |
| 443 | .sh 2 "linear current-controlled voltage sources" |
| 444 | .sp 0.2i |
| 445 | .b "General form:" |
| 446 | .(l |
| 447 | hxxxxxxx n+ n- vnam value |
| 448 | .)l |
| 449 | .sp 0.2i |
| 450 | .b "Examples:" |
| 451 | .(l |
| 452 | hx 5 17 vz 0.5k |
| 453 | .)l |
| 454 | .pp |
| 455 | N+ and n- are the positive and negative nodes, respectively. Vnam is the |
| 456 | name of a voltage source through which the controlling current flows. The |
| 457 | direction of positive controlling current flow is from the positive node, |
| 458 | through the source, to the negative node of vnam. Value is the transresistance |
| 459 | (in ohms). |
| 460 | .sh 2 "independent sources" |
| 461 | .sp 0.2i |
| 462 | .b "General form:" |
| 463 | .(l |
| 464 | vxxxxxxx n+ n- <<dc> dc/tran value> <ac <acmag <acphase>>> |
| 465 | .)l |
| 466 | iyyyyyyy n+ n- <<dc> dc/tran value> <ac <acmag <acphase>>> |
| 467 | .sp 0.2i |
| 468 | .b "Examples:" |
| 469 | .(l |
| 470 | vcc 10 0 dc 6 |
| 471 | vin 13 2 0.001 ac 1 sin(0 1 1meg) |
| 472 | isrc 23 21 ac 0.333 45.0 sffm(0 1 10k 5 1k) |
| 473 | vmeas 12 9 |
| 474 | .)l |
| 475 | .pp |
| 476 | N+ and n- are the positive and negative nodes, respectively. Note that |
| 477 | voltage sources need not be grounded. Positive current is assumed to flow from |
| 478 | positive node, through the source, to the negative node. |
| 479 | A current sources of positive value, will force current to flow out of |
| 480 | the n+ node, through the source, and into the n- node. |
| 481 | Voltage sources, in addition to being |
| 482 | used for circuit excitation, are the 'ammeters' for spice, |
| 483 | that is, zero valued voltage sources may be inserted into the circuit for the pu |
| 484 | of measuring current. They will, of course, have no effect on circuit |
| 485 | operation since they represent short-circuits. |
| 486 | .sp 0.2i |
| 487 | .pp |
| 488 | Dc/tran is the dc and transient analysis value of the source. If the |
| 489 | source value is zero both for dc and transient analyses, this value may be |
| 490 | omitted. If the source value is time-invariant (e.g., a power supply), then |
| 491 | the value may optionally be preceded by the letters dc. |
| 492 | .sp 0.2i |
| 493 | .pp |
| 494 | Acmag is the ac magnitude and acphase is the ac phase. The source is set |
| 495 | to this value in the ac analysis. If acmag is omitted following the keyword |
| 496 | ac, a value of unity is assumed. If acphase is omitted, a value of zero is |
| 497 | assumed. If the source is not an ac small-signal input, the keyword ac and the |
| 498 | ac values are omitted. |
| 499 | .sp 0.2i |
| 500 | .pp |
| 501 | Any independent source can be assigned a time-dependent value for |
| 502 | transient analysis. If a source is assigned adependent value, the time- |
| 503 | time-zero value is used for dc analysis. There are five independent source |
| 504 | functions: pulse, exponential, sinusoidal, piece-wise linear, and single-freque |
| 505 | fm. If parameters other than source values are omitted or set to zero, the |
| 506 | default values shown will be assumed. (tstep is the printing increment and |
| 507 | tstop is the final time (see the .tran card for explanation)). |
| 508 | .sp 0.2i |
| 509 | 1. Pulse pulse(v1 v2 td tr tf pw per) |
| 510 | .sp 0.2i |
| 511 | .b "Examples:" |
| 512 | .(l |
| 513 | vin 3 0 pulse(-1 1 2ns 2ns 2ns 50ns 100ns) |
| 514 | .)l |
| 515 | .TS |
| 516 | center; |
| 517 | l l l. |
| 518 | parameters default values units |
| 519 | .sp 0.2i |
| 520 | v1 (initial value) volts or amps |
| 521 | v2 (pulsed value) volts or amps |
| 522 | td (delay time) 0.0 seconds |
| 523 | tr (rise time) tstep seconds |
| 524 | tf (fall time) tstep seconds |
| 525 | pw (pulse width) tstop seconds |
| 526 | per (period) tstop seconds |
| 527 | .TE |
| 528 | .pp |
| 529 | A single pulse so specified is described by the following table: |
| 530 | .sp 0.2i |
| 531 | .TS |
| 532 | center; |
| 533 | l l. |
| 534 | time value |
| 535 | .sp 0.2i |
| 536 | 0 v1 |
| 537 | td v1 |
| 538 | td+tr v2 |
| 539 | td+tr+pw v2 |
| 540 | td+tr+pw+tf v1 |
| 541 | tstop v1 |
| 542 | .TE |
| 543 | .sp 0.1i |
| 544 | Intermediate points are determined by linear interpolation. |
| 545 | .sp 0.1i |
| 546 | 2. Sinusoidal sin(vo va freq td theta) |
| 547 | .sp 0.2i |
| 548 | .b "Examples:" |
| 549 | .(l |
| 550 | vin 3 0 sin(0 1 100meg 1ns 1e10) |
| 551 | .)l |
| 552 | .sp 0.2i |
| 553 | .TS |
| 554 | center; |
| 555 | l l l. |
| 556 | parameters default value units |
| 557 | .sp 0.2i |
| 558 | vo (offset) volts or amps |
| 559 | va (amplitude) volts or amps |
| 560 | freq (frequency) 1/tstop hz |
| 561 | td (delay) 0.0 seconds |
| 562 | theta (damping factor) 0.0 1/seconds |
| 563 | .TE |
| 564 | .pp |
| 565 | The shape of the waveform is described by the following table: |
| 566 | .TS |
| 567 | center; |
| 568 | l l. |
| 569 | .sp 0.2i |
| 570 | time value |
| 571 | .sp 0.2i |
| 572 | 0 to td vo |
| 573 | td to tstop vo + va*exp(-(time-td)*theta)*sine(twopi*freq*(time-td)) |
| 574 | .TE |
| 575 | .sp 0.2i |
| 576 | .bp |
| 577 | 3. Exponential exp(v1 v2 td1 tau1 td2 tau2) |
| 578 | .sp 0.2i |
| 579 | .b "Examples:" |
| 580 | .(l |
| 581 | vin 3 0 exp(-4 -1 2ns 30ns 60ns 40ns) |
| 582 | .)l |
| 583 | .sp 0.2i |
| 584 | .TS |
| 585 | center; |
| 586 | l l. |
| 587 | parameters default values units |
| 588 | .sp 0.2i |
| 589 | v1 (initial value) volts or amps |
| 590 | v2 (pulsed value) volts or amps |
| 591 | td1 (rise delay time) 0.0 seconds |
| 592 | tau1 (rise time constant) tstep seconds |
| 593 | td2 (fall delay time) td1+tstep seconds |
| 594 | tau2 (fall time constant) tstep seconds |
| 595 | .TE |
| 596 | .pp |
| 597 | The shape of the waveform is described by the following table: |
| 598 | .sp 0.2i |
| 599 | .TS |
| 600 | center; |
| 601 | l l. |
| 602 | time value |
| 603 | .sp 0.2i |
| 604 | 0 to td1 v1 |
| 605 | td1 to td2 v1+(v2-v1)*(1-exp(-(time-td1)/tau1)) |
| 606 | td2 to tstop v1+(v2-v1)*(1-exp(-(time-td1)/tau1)) |
| 607 | +(v1-v2)*(1-exp(-(time-td2)/tau2)) |
| 608 | .TE |
| 609 | .sp 0.2i |
| 610 | 4. Piece-wise linear |
| 611 | .sp 0.2i |
| 612 | pwl(t1 v1 <t2 v2 t3 v3 t4 v4 ...>) |
| 613 | .sp 0.2i |
| 614 | .b "Examples:" |
| 615 | .(l |
| 616 | vclock 7 5 pwl(0 -7 10ns -7 11ns -3 17ns -3 18ns -7 50ns -7) |
| 617 | .)l |
| 618 | .sp 0.2i |
| 619 | .TS |
| 620 | center; |
| 621 | l l. |
| 622 | parameters default values |
| 623 | .TE |
| 624 | .(l |
| 625 | Each pair of values (ti, vi) specifies that the value of the source is vi |
| 626 | (in volts or amps) at time=ti. The value of the source at intermediate values |
| 627 | of time is determined by using linear interpolation on the input values. |
| 628 | .)l |
| 629 | .sp 0.2i |
| 630 | .bp |
| 631 | 5. Single-frequency fm |
| 632 | .sp 0.2i |
| 633 | sffm(vo va fc mdi fs) |
| 634 | .sp 0.2i |
| 635 | .b "Examples:" |
| 636 | .(l |
| 637 | v1 12 0 sffm(0 1m 20k 5 1k) |
| 638 | .)l |
| 639 | .sp 0.2i |
| 640 | .TS |
| 641 | center; |
| 642 | l l l. |
| 643 | parameters default values units |
| 644 | .sp 0.2i |
| 645 | vo (offset) volts or amps |
| 646 | va (amplitude) volts or amps |
| 647 | fc (carrier frequency) 1/tstop hz |
| 648 | mdi (modulation index) |
| 649 | fs (signal frequency) 1/tstop hz |
| 650 | .TE |
| 651 | .pp |
| 652 | The shape of the waveform is described by the following equation: |
| 653 | .(l |
| 654 | value = vo + va*sine((twopi*fc*time) + mdi*sine(twopi*fs*time)) |
| 655 | .)l |
| 656 | .bp |
| 657 | .sh 1 "SEMICONDUCTOR DEVICES" |
| 658 | .pp |
| 659 | The elements that have been described to this point typically require only |
| 660 | a few parameter values to specify completely the electrical characteristics of |
| 661 | the element. However, the models for the four semiconductor devices that are |
| 662 | included in the spice program require many parameter values. Moreover, many |
| 663 | devices in a circuit often are defined by the same set of device model |
| 664 | parameters. For these reasons, a set of device model parameters is defined on a |
| 665 | separate .model card and assigned a unique model name. The device element |
| 666 | cards in spice then reference the model name. This scheme alleviates the need |
| 667 | to specify all of the model parameters on each device element card. |
| 668 | .pp |
| 669 | Each device element card contains the device name, the nodes to which the |
| 670 | device is connected, and the device model name. In addition, two optional |
| 671 | parameters may be specified for each device: an area factor, and an initial |
| 672 | condition. |
| 673 | .pp |
| 674 | The area factor determines the number of equivalent parallel devices of a |
| 675 | specified model. The affected parameters are marked with an asterisk under the |
| 676 | heading 'area' in the model descriptions below. |
| 677 | .pp |
| 678 | Two different forms of initial conditions may be specified for devices. |
| 679 | The first form is included to improve the dc convergence for circuits that |
| 680 | contain more than one stable state. If a device is specified off, the dc |
| 681 | operating point is determined with the terminal voltages for that device set to |
| 682 | zero. After convergence is obtained, the program continues to iterate to |
| 683 | obtain the exact value for the terminal voltages. If a circuit has more than |
| 684 | one dc stable state, the off option can be used to force the solution to |
| 685 | correspond to a desired state. If a device is specified off when in reality |
| 686 | the device is conducting, the program will still obtain the correct solution |
| 687 | (assuming the solutions converge) but more iterations will be required since |
| 688 | the program must independently converge to two separate solutions. |
| 689 | The .nodeset card serves a similar purpose as the 'off' option. The .nodeset |
| 690 | option is easier to apply and is the preferred means to aid convergence. |
| 691 | .pp |
| 692 | The second form of initial conditions are specified for use with |
| 693 | the transient analysis. These are true 'initial conditions' as opposed |
| 694 | to the convergence aids above. See the description of the .ic card and |
| 695 | the .tran card for a detailed explanation of initial conditions. |
| 696 | .sh 2 "junction diodes" |
| 697 | .sp 0.2i |
| 698 | .b "General form:" |
| 699 | .(l |
| 700 | dxxxxxxx n+ n- mname <area> <off> <ic=vd> |
| 701 | .)l |
| 702 | .sp 0.2i |
| 703 | .b "Examples:" |
| 704 | .(l |
| 705 | dbridge 2 10 diode1 |
| 706 | dclmp 3 7 dmod 3.0 ic=0.2 |
| 707 | .)l |
| 708 | .pp |
| 709 | N+ and n- are the positive and negative nodes, respectively. Mname is the |
| 710 | model name, area is the area factor, and off indicates an (optional) starting |
| 711 | condition on the device for dc analysis. If the area factor is omitted, a |
| 712 | value of 1.0 is assumed. The (optional) initial condition specification using |
| 713 | ic=vd is intended for use with the uic option on the .tran card, when a |
| 714 | transient analysis is desired starting from other than the quiescent operating |
| 715 | point. |
| 716 | .sp 0.2i |
| 717 | .sh 2 "bipolar junction transistors (bjt's)" |
| 718 | .sp 0.2i |
| 719 | .b "General form:" |
| 720 | .(l |
| 721 | qxxxxxxx nc nb ne <ns> mname <area> <off> <ic=vbe,vce> |
| 722 | .)l |
| 723 | .sp 0.2i |
| 724 | .b "Examples:" |
| 725 | .(l |
| 726 | q23 10 24 13 qmod ic=0.6,5.0 |
| 727 | q50a 11 26 4 20 mod1 |
| 728 | .)l |
| 729 | .pp |
| 730 | Nc, nb, and ne are the collector, base, and emitter nodes, respectively. |
| 731 | Ns is the (optional) substrate node. If unspecified, ground is used. |
| 732 | mname is the model name, area is the area factor, and off indicates an |
| 733 | (optional) initial condition on the device for the dc analysis. If the area |
| 734 | factor is omitted, a value of 1.0 is assumed. The (optional) initial condition |
| 735 | specification using ic=vbe,vce is intended for use with the uic option on |
| 736 | the .tran card, when a transient analysis is desired starting from other than th |
| 737 | quiescent operating point. See the '.ic' card description for a better way to |
| 738 | set transient initial conditions. |
| 739 | .sp 0.2i |
| 740 | .sh 2 "junction field-effect transistors (jfet's)" |
| 741 | .sp 0.2i |
| 742 | .b "General form:" |
| 743 | .(l |
| 744 | jxxxxxxx nd ng ns mname <area> <off> <ic=vds,vgs> |
| 745 | .)l |
| 746 | .sp 0.2i |
| 747 | .b "Examples:" |
| 748 | .(l |
| 749 | j1 7 2 3 jm1 off |
| 750 | .)l |
| 751 | .pp |
| 752 | Nd, ng, and ns are the drain, gate, and source nodes, respectively. Mname |
| 753 | is the model name, area is the area factor, and off indicates an (optional) |
| 754 | initial condition on the device for dc analysis. If the area factor is |
| 755 | omitted, a value of 1.0 is assumed. The (optional) initial condition specification, |
| 756 | using ic=vds,vgs is intended for use with the uic option on the .tran card, |
| 757 | when a transient analysis is desired starting from other than the quiescent |
| 758 | operating point (see the .ic card for a better way to set initial conditions). |
| 759 | .sp 0.2i |
| 760 | .sh 2 "mosfets" |
| 761 | .sp 0.2i |
| 762 | .b "General form:" |
| 763 | .(l |
| 764 | mxxxxxxx nd ng ns nb mname <l=val> <w=val> <ad=val> <as=val> |
| 765 | + <rd=val> <rs=val> <off> <ic=vds,vgs,vbs> |
| 766 | .)l |
| 767 | .sp 0.2i |
| 768 | .b "Examples:" |
| 769 | .(l |
| 770 | m1 24 2 0 20 type1 |
| 771 | m31 2 17 6 10 modm l=5u w=2u |
| 772 | m31 2 16 6 10 modm 5u 2u |
| 773 | m1 2 9 3 0 mod1 l=10u w=5u ad=2p as=2p |
| 774 | m1 2 9 3 0 mod1 10u 5u 2p 2p |
| 775 | .)l |
| 776 | Nd, ng, ns, and nb are the drain, gate, source, and bulk (substrate) |
| 777 | nodes, respectively. Mname is the model name. L and w are the channel length |
| 778 | and width, in meters. Ad and as are the areas of the drain and source |
| 779 | diffusions, in sq-meters. Note that the suffix 'u' specifies microns (10**-6 m) |
| 780 | and 'p' sq-microns (10**-12 sq-m). If any of l, w, ad, or as are not specified, |
| 781 | default values are used. The user may specify the values to be used for |
| 782 | these default parameters on the .option card. The use of defaults simplifies |
| 783 | input deck preparation, as well as the editing required if devices geometries |
| 784 | are to be changed. Off indicates an (optional) initial condition |
| 785 | on the device for dc analysis. The (optional) initial condition |
| 786 | specification using ic=vds,vgs,vbs is intended for use with the uic option |
| 787 | on the .tran card, when a transient analysis is desired starting from other |
| 788 | than the quiescent operating point. See the .ic card for a better and |
| 789 | more convenient way to specify transient initial conditions. |
| 790 | .bp |
| 791 | .sp 0.2i |
| 792 | .sh 2 ".model card" |
| 793 | .sp 0.2i |
| 794 | .b "General form:" |
| 795 | .(l |
| 796 | .model mname type(pname1=pval1 pname2=pval2 ... ) |
| 797 | .)l |
| 798 | .sp 0.2i |
| 799 | .b "Examples:" |
| 800 | .(l |
| 801 | .model mod1 npn bf=50 js=1e-13 vbf=50 |
| 802 | .)l |
| 803 | .pp |
| 804 | The .model card specifies a set of model parameters that will be used by |
| 805 | one or more devices. Mname is the model name, and type is one of the following |
| 806 | seven types: |
| 807 | .TS |
| 808 | center; |
| 809 | l l. |
| 810 | npn npn bjt model |
| 811 | pnp pnp bjt model |
| 812 | d diode model |
| 813 | njf n-channel jfet model |
| 814 | pjf p-channel jfet model |
| 815 | nmos n-channel mosfet model |
| 816 | pmos p-channel mosfet model |
| 817 | .TE |
| 818 | .pp |
| 819 | Parameter values are defined by appending the parameter name, as given |
| 820 | below for each model type, followed by an equal sign and the parameter value. |
| 821 | Model parameters that are not given a value are assigned the default values |
| 822 | given below for each model type. |
| 823 | .sp 0.2i |
| 824 | .sh 2 "diode model" |
| 825 | .pp |
| 826 | The dc characteristics of the diode are determined by the parameters is |
| 827 | and n. An ohmic resistance, rs, is included. Charge storage effects are |
| 828 | modeled by a transit time, tt, and a nonlinear depletion layer capacitance |
| 829 | which is determined by the parameters cjo, pb, and m. The temperature |
| 830 | dependence of the saturation current is defined by the parameters eg, the energy |
| 831 | and pt, the saturation current temperature exponent. Reverse breakdown is |
| 832 | modeled by an exponential increase in the reverse diode current and is |
| 833 | determined by the parameters bv and ibv (both of which are positive numbers). |
| 834 | .sp 0.2i |
| 835 | .TS |
| 836 | center; |
| 837 | l l l l l l. |
| 838 | area name parameter default example |
| 839 | .sp 0.2i |
| 840 | 1 * is saturation current 1.0e-14 1.0e-14 |
| 841 | 2 * rs ohmic resistance 0 10 |
| 842 | 3 n emission coefficient 1 1.0 |
| 843 | 4 tt transit-time 0 0.1ns |
| 844 | 5 * cjo zero-bias junction capacitance 0 2pf |
| 845 | 6 pb junction potential 1 0.6 |
| 846 | 7 m grading coefficient 0.5 0.5 |
| 847 | 8 eg activation energy 1.11 1.11 si |
| 848 | 0.69 sbd |
| 849 | 0.67 ge |
| 850 | 9 pt saturation-current temp. exp 3.0 3.0 jn |
| 851 | 2.0 sbd |
| 852 | 10 kf flicker noise coefficient 0 |
| 853 | 11 af flicker noise exponent 1 |
| 854 | 12 fc coefficient for forward-bias 0.5 |
| 855 | depletion capacitance formula |
| 856 | 13 bv reverse breakdown voltage infinite 40.0 |
| 857 | 14 ibv current at breakdown voltage 1.0e-3 |
| 858 | .TE |
| 859 | .sh 2 "bjt models (both npn and pnp)" |
| 860 | .pp |
| 861 | The bipolar junction transistor model in spice is an adaptation of |
| 862 | the integral charge control model of Gummel and Poon. This modified |
| 863 | Gummel-Poon model extends the original model to include several effects |
| 864 | at high bias levels. The model will automatically simplify to the simpler |
| 865 | Ebers-Moll model when certain parameters are not specified. To permit |
| 866 | one to use model parameters from earlier versions of spice, many |
| 867 | of the model parameters can be called by two names. The parameter names |
| 868 | used in the modified Gummel-Poon model have been chosen to be more easily |
| 869 | understood by the program user, and to better reflect both physical and |
| 870 | circuit design thinking. The dc model is defined by the parameters bf, |
| 871 | jbf, jle, and nle which determine the forward current gain characteristics, |
| 872 | br, jbr, jlc, and nlc which determine the reverse current gain characteristics, |
| 873 | vbf and vbr, which determine the output conductance for forward and reverse |
| 874 | regions, and the saturation current, js. Three ohmic resistances rb, rc, and |
| 875 | re are included, where rb can be high current dependent. Base charge storage |
| 876 | is modeled by forward and reverse transit times, tf and tr the forward transit |
| 877 | time being bias dependent if desired, and nonlinear depletion layer |
| 878 | capacitances which are determined by cje, vje, and mje for the b-e junction and |
| 879 | cjc, vjc, and mjc for the b-c junction. A depletion formulation is used for |
| 880 | the substrate capacitance described by cjs, vjs, and mjs. The temperature |
| 881 | dependence of saturation current, js, is determined by the energy-gap, eg, |
| 882 | and the saturation current temperature exponent, pt. Base current temperature |
| 883 | dependence is modeled by the temperature exponent for beta, tb. |
| 884 | .sp 0.2i |
| 885 | .TS |
| 886 | center; |
| 887 | l l l l. |
| 888 | name parameter units default |
| 889 | .sp 0.2i |
| 890 | js transport saturation current amps 1.0e-16 |
| 891 | bf ideal maximum forward beta amp/amp 100 |
| 892 | nf forward current emission coefficient - 1.0 |
| 893 | vbf forward early voltage volts infinite |
| 894 | jbf corner for forward beta high current roll-off amps infinite |
| 895 | jle base-emitter leakage saturation current amps 0 |
| 896 | nle base-emitter leakage emission coefficient - 1.5 |
| 897 | br ideal maximum reverse beta amp/amp 1.0 |
| 898 | nr reverse current emission coefficient - 1.0 |
| 899 | vbr reverse early voltage volts infinite |
| 900 | jbr corner for reverse beta high current roll-off amps infinite |
| 901 | jlc base-collector leakage saturation current amps 0 |
| 902 | nlc base-collector leakage emission coefficient - 2.0 |
| 903 | rb zero bias base resistance ohms 0 |
| 904 | jrb current where base resistance falls halfway to amps infinite |
| 905 | its minimum value |
| 906 | rbm minimum base resistance at high currents ohms rb |
| 907 | re emitter resistance ohms 0 |
| 908 | rc collector resistance ohms 0 |
| 909 | cje base-emitter zero bias depletion capacitance farads 0 |
| 910 | vje base-emitter built-in potential volts .75 |
| 911 | mje base-emitter junction exponential factor - .33 |
| 912 | tf ideal forward transit time sec 0 |
| 913 | xtf coefficient for bias dependence of tf - 0 |
| 914 | vtf voltage describing vbc dependence of tf volts infinite |
| 915 | jtf high-current parameter for effect on tf amps 0 |
| 916 | ptf excess phase at freq=1.0/(tf*2pi) hz degrees 0 |
| 917 | cjc base-collector zero bias depletion capacitance farads 0 |
| 918 | vjc base-collector built-in potential volts .75 |
| 919 | mjc base-collector junction exponential factor - .33 |
| 920 | cdis fraction of base-collector depletion - 1.0 |
| 921 | capacitance connected to internal base node |
| 922 | tr ideal reverse transit time sec 0 |
| 923 | cjs zero bias substrate capacitance farads 0 |
| 924 | vjs substrate junction built-in potential volts .75 |
| 925 | mjs substrate junction exponential factor - 0 |
| 926 | tb forward and reverse beta temperature exponent - 0 |
| 927 | eg energy-gap for temperature effect on js ev 1.11 |
| 928 | pt temperature exponent for effect on js - 3 |
| 929 | kf flicker-noise coefficient - 0 |
| 930 | af flicker-noise exponent - 1 |
| 931 | fc coefficient for forward-bias depletion - .5 |
| 932 | capacitance formula |
| 933 | .TE |
| 934 | .sp 0.2i |
| 935 | .sh 2 "jfet models (both n and p channel)" |
| 936 | .sp 0.2i |
| 937 | .pp |
| 938 | The jfet model is derived from the fet model of Shichman and Hodges. The |
| 939 | dc characteristics are defined by the parameters vto and beta, which determine |
| 940 | the variation of drain current with gate voltage, lambda, which determines the |
| 941 | output conductance, and is, the saturation current of the two gate junctions. |
| 942 | Two ohmic resistances, rd and rs, are included. Charge storage is modeled by |
| 943 | nonlinear depletion layer capacitances for both gate junctions which vary as |
| 944 | the -1/2 power of junction voltage and are defined by the parameters cgs, cgd, |
| 945 | and pb. |
| 946 | .sp 0.2i |
| 947 | .TS |
| 948 | center; |
| 949 | l l l l l l. |
| 950 | area name parameter default example |
| 951 | .sp 0.2i |
| 952 | 1 vto threshold voltage -2.0 -2.0 |
| 953 | 2 * beta transconductance parameter 1.0e-4 1.0e-3 |
| 954 | 3 lambda channel length modulation parameter 0 1.0e-4 |
| 955 | 4 * rd drain ohmic resistance 0 100 |
| 956 | 5 * rs source ohmic resistance 0 100 |
| 957 | 6 * cgs zero-bias g-s junction capacitance 0 5pf |
| 958 | 7 * cgd zero-bias g-d junction capacitance 0 1pf |
| 959 | 8 pb gate junction potential 1 0.6 |
| 960 | 9 * is gate junction saturation current 1.0e-14 1.0e-14 |
| 961 | 10 kf flicker noise coefficient 0 |
| 962 | 11 af flicker noise exponent 1 |
| 963 | 12 fc coefficient for forward-bias 0.5 |
| 964 | depletion capacitance formula |
| 965 | .TE |
| 966 | .sp 0.2i |
| 967 | .sh 2 "mosfet models (both n and p channel)" |
| 968 | .sp 0.2i |
| 969 | The dc mosfet equations |
| 970 | are determined by the parameters vto, kp, gamma, lambda, and phi. These |
| 971 | parameters may be specified by the user, or they will be computed from |
| 972 | values specified for nsub, tox, nss, nfs, ngate, tps, uo, ucrit, uexp, and |
| 973 | utra. Vto is positive (negative) for enhancement mode and negative |
| 974 | (posiive) for depletion mode n-channel (p-channel) devices. Charge storage is |
| 975 | modeled by three constant capacitors, cgs, cgd, and cgb, by the nonlinear oxide |
| 976 | gate capacitance which is distributed among the gate-source, gate-drain, and |
| 977 | bulk regions using the formulation of J.E. Meyer, and by the nonlinear |
| 978 | depletion-layer capacitances for both substrate junctions which vary as the -1/2 |
| 979 | power of junction voltage and are determined by the parameters cbd, cbs, and |
| 980 | pb. |
| 981 | .sp 0.2i |
| 982 | .TS |
| 983 | center; |
| 984 | l l l l l l. |
| 985 | name parameter default example units |
| 986 | .sp 0.2i |
| 987 | 1 vto zero-bias threshold voltage 0.0 1.0 v |
| 988 | 2 kp intrinsic transconductance parameter 2.417e-5 3.1e-5 a/v**2 |
| 989 | 3 gamma bulk threshold parameter 0.0 0.37 v**(1/2) |
| 990 | 4 phi surface potential at strong inversion 0.6 0.65 v |
| 991 | 5 lambda channel-length modulation parameter 0.0 1.0e-7 meters/v |
| 992 | 6 rd drain ohmic resistance 0.0 1.0 ohms |
| 993 | 7 rs source ohmic resistance 0.0 1.0 ohms |
| 994 | 8 cgs gate-source overlap capacitance |
| 995 | per meter channel width 0.0 4.0e-11 f/m |
| 996 | 9 cgd gate-drain overlap capacitance |
| 997 | per meter channel width 0.0 4.0e-11 f/m |
| 998 | 10 cgb gate-bulk overlap capacitance |
| 999 | per meter channel length 0.0 2.0e-10 f/m |
| 1000 | 11 cbd zero-bias b-d junction capacitance |
| 1001 | per sq-meter of junction area 0.0 2.0e-4 f/sq-m |
| 1002 | 12 cbs zero-bias b-s junction capacitance |
| 1003 | per sq-meter of junction area 0.0 2.0e-4 f/sq-m |
| 1004 | 13 tox oxide thickness 1.0e-7 1.0e-7 meters |
| 1005 | 14 pb bulk junction potential 0.8 0.87 v |
| 1006 | 15 js bulk junction reverse saturation current |
| 1007 | per sq-meter of junction area 1.0e-4 1.0e-4 a/sq-m |
| 1008 | 16 nsub substrate doping 0.0 4.0e15 /cm**3 |
| 1009 | 17 nss surface state density 0.0 1.0e10 /cm**2 |
| 1010 | 18 nfs fast surface state density 0.0 1.0e10 /cm**2 |
| 1011 | 19 xj metallurgical junction depth 0.0 1.0e-6 meters |
| 1012 | 20 ld lateral diffusion (channel length is 0.0 0.8e-6 meters |
| 1013 | reduced such that leff=l-2*ld) |
| 1014 | 21 wd width reduction (channel width is 0.0 1.0e-6 meters |
| 1015 | reduced such that weff=w-2*wd) |
| 1016 | 22 ngate polysilicon gate doping al gate 1.0e20 /cm**3 |
| 1017 | 23 tps type of polysilicon: +1 opp to sub 1.0 |
| 1018 | -1 same as sub |
| 1019 | 24 uo surface mobility 700 600 cm**2/v-s |
| 1020 | 25 ucrit critical field for mobility 1.0e+4 1.0e+4 v/cm |
| 1021 | 26 uexp critical field exponent (mobility) 0.0 0.1 |
| 1022 | 27 utra transverse field coefficient (mobility) 0.0 0.3 |
| 1023 | 28 kf flicker noise coefficient 0.0 |
| 1024 | 29 af flicker noise exponent 1.0 |
| 1025 | 30 fc coefficient for forward-bias 0.5 |
| 1026 | depletion capacitance formula |
| 1027 | .TE |
| 1028 | .bp |
| 1029 | .sh 1 "SUBCIRCUITS" |
| 1030 | .pp |
| 1031 | A subcircuit that consists of spice elements can be defined and referenced |
| 1032 | in a fashion similar to device models. The subcircuit is defined in the input |
| 1033 | deck by a grouping of element cards; the program then automatically inserts |
| 1034 | the group of elements wherever the subcircuit is referenced. There is no limit |
| 1035 | on the size or complexity of subcircuits, and subcircuits may contain other |
| 1036 | subcircuits. An example of subcircuit usage is given in Appendix A. |
| 1037 | .sp 0.2i |
| 1038 | .sh 2 ".subckt card" |
| 1039 | .sp 0.2i |
| 1040 | .b "General form:" |
| 1041 | .(l |
| 1042 | .subckt subnam n1 <n2 n3 ...> |
| 1043 | .)l |
| 1044 | .b "Examples:" |
| 1045 | .(l |
| 1046 | .subckt opamp 1 2 3 4 |
| 1047 | .)l |
| 1048 | .pp |
| 1049 | A subcircuit definition is begun with a .subckt card. Subnam is the |
| 1050 | subcircuit name, and n1, n2, ... Are the external nodes, which cannot be zero. |
| 1051 | The group of element cards which immediately follow the .subckt card define the |
| 1052 | subcircuit. The last card in a subcircuit definition is the .ends card (see |
| 1053 | below). Control cards may not appear within a subcircuit definition; however, |
| 1054 | subcircuit definitions may contain anything else, including other subcircuit |
| 1055 | definitions, device models, and subcircuit calls (see below). Note that any |
| 1056 | device models or subcircuit definitions included as part of a subcircuit |
| 1057 | definition are strictly local (i.e., such models and definitions are not known |
| 1058 | outside the subcircuit definition). Also, any element nodes not included on |
| 1059 | the .subckt card are strictly local, with the exception of 0 (ground) which is |
| 1060 | always global. |
| 1061 | .sh 2 ".ends card" |
| 1062 | .sp 0.2i |
| 1063 | .b "General form:" |
| 1064 | .(l |
| 1065 | .ends <subnam> |
| 1066 | .)l |
| 1067 | .b "Examples:" |
| 1068 | .(l |
| 1069 | .ends opamp |
| 1070 | .)l |
| 1071 | .pp |
| 1072 | This card must be the last one for any subcircuit definition. The sub- |
| 1073 | circuit name, if included, indicates which subcircuit definition is being |
| 1074 | terminated; if omitted, all subcircuits being defined are terminated. The |
| 1075 | name is needed only when nested subcircuit definitions are being made. |
| 1076 | .sp 0.2i |
| 1077 | .sh 2 "subcircuit calls" |
| 1078 | .sp 0.2i |
| 1079 | .b "General form:" |
| 1080 | .(l |
| 1081 | xyyyyyyy n1 <n2 n3 ...> subnam |
| 1082 | .)l |
| 1083 | .sp 0.2i |
| 1084 | .b "Examples:" |
| 1085 | .(l |
| 1086 | x1 2 4 17 3 1 multi |
| 1087 | .)l |
| 1088 | .pp |
| 1089 | Subcircuits are used in spice by specifying pseudo-elements beginning with |
| 1090 | the letter x, followed by the circuit nodes to be used in expanding the sub- |
| 1091 | circuit. |
| 1092 | .bp |
| 1093 | .sh 1 "CONTROL CARDS" |
| 1094 | .sp 0.2i |
| 1095 | .sh 2 ".temp card" |
| 1096 | .sp 0.2i |
| 1097 | .b "General form:" |
| 1098 | .(l |
| 1099 | .temp t1 <t2 <t3 ...>> |
| 1100 | .)l |
| 1101 | .b "Examples:" |
| 1102 | .(l |
| 1103 | .temp -55.0 25.0 125.0 |
| 1104 | .)l |
| 1105 | .pp |
| 1106 | This card specifies the temperatures at which the circuit is to be |
| 1107 | simulated. T1, t2, ... Are the different temperatures, in degrees c. Temperatu |
| 1108 | less than -223.0 deg c are ignored. Model data is specified at tnom degrees |
| 1109 | (see the .option card for tnom); if the .temp card is omitted, the simulation |
| 1110 | also will be performed at a temperature equal to tnom. |
| 1111 | .sp 0.2i |
| 1112 | .sh 2 ".width card" |
| 1113 | .sp 0.2i |
| 1114 | .b "General form:" |
| 1115 | .(l |
| 1116 | .width in=colnum out=colnum |
| 1117 | .)l |
| 1118 | .sp 0.2i |
| 1119 | .b "Examples:" |
| 1120 | .(l |
| 1121 | .width in=72 out=133 |
| 1122 | .)l |
| 1123 | .pp |
| 1124 | Colnum is the last column read from each line of input; the setting takes |
| 1125 | effect with the next line read. The default value for colnum is 80. |
| 1126 | The out parameter specifies the output print width. Permissible values for |
| 1127 | the output print width are 80 and 133. |
| 1128 | .sp 0.2i |
| 1129 | .sh 2 ".options card" |
| 1130 | .sp 0.2i |
| 1131 | .b "General form:" |
| 1132 | .(l |
| 1133 | .options opt1 opt2 ... (or opt=optval ...) |
| 1134 | .)l |
| 1135 | .b "Examples:" |
| 1136 | .(l |
| 1137 | .options noacct nolist nonode |
| 1138 | .)l |
| 1139 | .pp |
| 1140 | This card allows the user to reset program control and user options for |
| 1141 | specific simulation purposes. Any combination of the following options may be |
| 1142 | included, in any order. 'x' (below) represents some positive number. |
| 1143 | .TS |
| 1144 | center; |
| 1145 | l l. |
| 1146 | option effect |
| 1147 | .sp 0.2i |
| 1148 | noacct supresses the listing of accounting and run time |
| 1149 | statistics. |
| 1150 | nolist supresses the summary listing of input data. |
| 1151 | nomod suppresses the printout of the model parameters. |
| 1152 | nopage suppresses page ejects |
| 1153 | nonode supresses the printing of the node table. |
| 1154 | opts causes the option values to be printed. |
| 1155 | gmin=x resets the value of gmin, the minimum conductance |
| 1156 | allowed by the program. The default value is 1.0e-12. |
| 1157 | reltol=x resets the relative error tolerance of the program. The |
| 1158 | default value is 0.001 (0.1 percent). |
| 1159 | abstol=x resets the absolute current error tolerance of the |
| 1160 | program. The default value is 1 picoamp. |
| 1161 | vntol=x resets the absolute voltage error tolerance of the |
| 1162 | program. The default value is 1 microvolt. |
| 1163 | trtol=x resets the transient error tolerance. The default value |
| 1164 | is 7.0. This parameter is an estimate of the factor by |
| 1165 | which spice overestimates the actual truncation error. |
| 1166 | chgtol=x resets the charge tolerance of the program. The default |
| 1167 | value is 1.0e-14. |
| 1168 | numdgt=x resets the number of significant digits printed for |
| 1169 | output variable values. X must satisfy the relation |
| 1170 | 0 < x < 8. The default value is 4. Note: this option is |
| 1171 | independent of the error tolerance used by spice (i.e., if |
| 1172 | the values of options reltol, abstol, etc. Are not changed |
| 1173 | then one may be printing numerical 'noise' for numdgt > 4. |
| 1174 | tnom=x resets the nominal temperature. The default value is |
| 1175 | 25 deg c (298 deg k). |
| 1176 | itl1=x resets the dc iteration limit. The default is 100. |
| 1177 | itl2=x resets the dc transfer curve iteration limit. The |
| 1178 | default is 50. |
| 1179 | itl3=x resets the lower transient analysis iteration limit. |
| 1180 | the default value is 4. |
| 1181 | itl4=x resets the transient analysis timepoint iteration limit. |
| 1182 | the default is 10. |
| 1183 | itl5=x resets the transient analysis total iteration limit. |
| 1184 | the default is 5000. Set itl5=0 to omit this test. |
| 1185 | cptime=x the maximum cpu-time in seconds allowed for this job. |
| 1186 | limtim=x resets the amount of cpu time reserved by spice for |
| 1187 | generating plots should a cpu time-limit cause job |
| 1188 | termination. The default value is 2 (seconds). |
| 1189 | limpts=x resets the total number of points that can be printed |
| 1190 | or plotted in a dc, ac, or transient analysis. The |
| 1191 | default value is 201. |
| 1192 | lvlcod=x if x is 2 (two), then machine code for the matrix |
| 1193 | solution will be generated. Otherwise, no machine code is |
| 1194 | generated. The default value is 2. Applies only to cdc |
| 1195 | computers. |
| 1196 | lvltim=x if x is 1 (one), the iteration timestep control is used. |
| 1197 | if x is 2 (two), the truncation-error timestep is used. |
| 1198 | the default value is 1. If method=Gear and maxord>2 then |
| 1199 | lvltim is set to 2 by spice. |
| 1200 | method=name sets the numerical integration method used by spice. |
| 1201 | Possible names are Gear or trapezoidal. The default is |
| 1202 | trapezoidal. |
| 1203 | maxord=x sets the maximum order for the integration method if |
| 1204 | Gear's variable-order method is used. X must be between |
| 1205 | 2 and 6. The default value is 2. |
| 1206 | defl=x sets the default value for mos channel length. |
| 1207 | defw=x sets the default value for mos channel width. |
| 1208 | defad=x sets the default value for mos drain diffusion area. |
| 1209 | defas=x sets the default value for mos source diffusion area. |
| 1210 | .TE |
| 1211 | .sp 0.2i |
| 1212 | .sh 2 ".op card" |
| 1213 | .sp 0.2i |
| 1214 | .b "General form:" |
| 1215 | .(l |
| 1216 | .op |
| 1217 | .)l |
| 1218 | .sp 0.2i |
| 1219 | .pp |
| 1220 | The inclusion of this card in an input deck will force spice to determine |
| 1221 | the dc operating point of the circuit with inductors shorted and capacitors |
| 1222 | opened. Note: a dc analysis is automatically performed prior to a transient |
| 1223 | analysis to determine the transient initial conditions, and prior to an ac |
| 1224 | small-signal analysis to determine the linearized, small-signal models for |
| 1225 | nonlinear devices. |
| 1226 | .pp |
| 1227 | Spice performs a dc operating point analysis if no other analyses are |
| 1228 | requested. |
| 1229 | .sp 0.2i |
| 1230 | .sh 2 ".dc card" |
| 1231 | .sp 0.2i |
| 1232 | .b "General form:" |
| 1233 | .(l |
| 1234 | .dc srcnam vstart vstop vincr [src2 start2 stop2 incr2] |
| 1235 | .)l |
| 1236 | .sp 0.2i |
| 1237 | .b "Examples:" |
| 1238 | .(l |
| 1239 | .dc vin 0.25 5.0 0.25 |
| 1240 | .dc vds 0 10 .5 vgs 0 5 1 |
| 1241 | .dc vce 0 10 .25 ib 0 10u 1u |
| 1242 | .)l |
| 1243 | .pp |
| 1244 | This card defines the dc transfer curve source and sweep limits. Srcnam |
| 1245 | is the name of an independent voltage or current source. Vstart, vstop, and |
| 1246 | vincr are the starting, final, and incrementing values respectively. The first |
| 1247 | example will cause the value of the voltage source vin to be swept from 0.25 |
| 1248 | volts to 5.0 volts in increments of 0.25 volts. A second source (src2) may |
| 1249 | optionally be specified with associated sweep parameters. In this case, |
| 1250 | the first source will be swept over its range for each value of the second |
| 1251 | source. This option can be useful for obtaining semiconductor device output |
| 1252 | characteristics. See the second example data deck in that section of the guide. |
| 1253 | .sp 0.2i |
| 1254 | .bp |
| 1255 | .sh 2 ".nodeset card" |
| 1256 | .sp 0.2i |
| 1257 | .b "General form:" |
| 1258 | .(l |
| 1259 | .nodeset v(nodnum)=val v(nodnum)=val ... |
| 1260 | .)l |
| 1261 | .b "Examples:" |
| 1262 | .(l |
| 1263 | .nodeset v(12)=4.5 v(4)=2.23 |
| 1264 | .)l |
| 1265 | .pp |
| 1266 | This card helps the program find the dc solution by making a preliminary |
| 1267 | pass with the specified nodes held to the given voltages. The restriction |
| 1268 | is then released and the iteration continues to the true solution. |
| 1269 | The .nodeset card may be necessary for convergence on bistable or astable |
| 1270 | circuits. In general, this card should not be necessary. |
| 1271 | .sp 0.2i |
| 1272 | .sh 2 ".ic card" |
| 1273 | .sp 0.2i |
| 1274 | .b "General form:" |
| 1275 | .(l |
| 1276 | .ic v(nodnum)=val v(nodnum)=val ... |
| 1277 | .)l |
| 1278 | .b "Examples:" |
| 1279 | .(l |
| 1280 | .ic v(11)=5 v(4)=-5 v(2)=2.2 |
| 1281 | .)l |
| 1282 | .pp |
| 1283 | This card is for setting transient initial conditions. It has two |
| 1284 | different interpretations, depending on whether the 'uic' parameter is |
| 1285 | specified on the '.tran' card. Also, one should not confuse this card with |
| 1286 | the '.nodeset' card. The '.nodeset' card is only to help dc convergence, |
| 1287 | and does not affect final bias solution (except for multi-stable circuits). |
| 1288 | The two interpretations of this card are as follows: |
| 1289 | .sp 0.2i |
| 1290 | 1. When the 'uic' parameter is specified on the '.tran' card, then |
| 1291 | .pp |
| 1292 | The node voltages specified on the '.ic' card are used to compute |
| 1293 | .pp |
| 1294 | The capacitor, diode, bjt, jfet, and mosfet initial conditions. |
| 1295 | .pp |
| 1296 | This is equivalent to specifying the 'ic=...' parameter on each |
| 1297 | .pp |
| 1298 | Device card, but is much more convenient. The 'ic=...' parameter |
| 1299 | .pp |
| 1300 | Can still be specified and will take precedence over the '.ic' |
| 1301 | .pp |
| 1302 | Values. Since no dc bias solution is computed before the transient |
| 1303 | .pp |
| 1304 | Analysis, one should take care to specify all dc source voltages |
| 1305 | .pp |
| 1306 | On the '.ic' card if they are to be used to compute device initial |
| 1307 | .pp |
| 1308 | Conditions. |
| 1309 | .sp 0.2i |
| 1310 | 2. When the 'uic' parameter is not specified on the '.tran' card, |
| 1311 | .pp |
| 1312 | The a dc bias solution will be computed before the transient analysis. |
| 1313 | .pp |
| 1314 | In this case, the node voltages specified on the '.ic' card will |
| 1315 | .pp |
| 1316 | Be forced to the desired initial values during the bias solution. |
| 1317 | .pp |
| 1318 | During transient analysis, the constraint on these node voltages |
| 1319 | is removed. |
| 1320 | .sp 0.2i |
| 1321 | .sh 2 ".tf card" |
| 1322 | .sp 0.2i |
| 1323 | .b "General form:" |
| 1324 | .(l |
| 1325 | .tf outvar insrc |
| 1326 | .)l |
| 1327 | .b "Examples:" |
| 1328 | .(l |
| 1329 | .tf v(5,3) vin |
| 1330 | .tf i(vload) vin |
| 1331 | .)l |
| 1332 | .pp |
| 1333 | This card defines the small-signal output and input for the dc small- |
| 1334 | signal analysis. Outvar is the small-signal output variable and insrc is the |
| 1335 | small-signal input source. If this card is included, spice will compute the |
| 1336 | dc small-signal value of the transfer function (outputinput), input |
| 1337 | resistance, and output resistance. For the first example, spice would compute t |
| 1338 | ratio of v(5,3) to vin, the small-signal input resistance at vin, and the |
| 1339 | small-signal output resistance measured across nodes 5 and 3. |
| 1340 | .sp 0.2i |
| 1341 | .sh 2 ".sens card" |
| 1342 | .sp 0.2i |
| 1343 | .b "General form:" |
| 1344 | .(l |
| 1345 | .sens ov1 <ov2 ... > |
| 1346 | .)l |
| 1347 | .b "Examples:" |
| 1348 | .(l |
| 1349 | .sens v(9) v(4,3) v(17) i(vcc) |
| 1350 | .)l |
| 1351 | .pp |
| 1352 | If a .sens card is included in the input deck, spice will determine the |
| 1353 | dc small-signal sensitivities of each specified output variable with respect to |
| 1354 | every circuit parameter. Note: for large circuits, large amounts of output |
| 1355 | can be generated. |
| 1356 | .sp 0.2i |
| 1357 | .sh 2 ".ac card" |
| 1358 | .sp 0.2i |
| 1359 | .b "General form:" |
| 1360 | .(l |
| 1361 | .ac dec nd fstart fstop |
| 1362 | .ac oct no fstart fstop |
| 1363 | .ac lin np fstart fstop |
| 1364 | .)l |
| 1365 | .b "Examples:" |
| 1366 | .(l |
| 1367 | .ac dec 10 1 10k |
| 1368 | .ac dec 10 1k 100meg |
| 1369 | .ac lin 100 1 100hz |
| 1370 | .)l |
| 1371 | .sp 0.2i |
| 1372 | .pp |
| 1373 | Dec stands for decade variation, and nd is the number of points per |
| 1374 | decade. Oct stands for octave variation, and no is the number of points per |
| 1375 | octave. Lin stands for linear variation, and np is the number of points. |
| 1376 | Fstart is the starting frequency, and fstop is the final frequency. If this |
| 1377 | card is included in the deck, spice will perform an ac analysis of the circuit |
| 1378 | over the specified frequency range. Note that in order for this analysis to be |
| 1379 | meaningful, at least one independent source must have been specified with an ac |
| 1380 | value. |
| 1381 | .sp 0.2i |
| 1382 | .sh 2 ".disto card" |
| 1383 | .sp 0.2i |
| 1384 | .b "General form:" |
| 1385 | .(l |
| 1386 | .disto rload <inter <skw2 <refpwr <spw2>>>> |
| 1387 | .)l |
| 1388 | .b "Examples:" |
| 1389 | .(l |
| 1390 | .disto rl 2 0.95 1.0e-3 0.75 |
| 1391 | .)l |
| 1392 | .pp |
| 1393 | This card controls whether spice will compute the distortion characteristic |
| 1394 | of the circuit in a small-signal mode as a part of the ac small-signal |
| 1395 | sinusoidal steady-state analysis. The analysis is performed assuming that |
| 1396 | one or two signal frequencies are imposed at the input; let the two frequencies |
| 1397 | be f1 (the nominal analysis frequency) and f2 (=skw2*f1). The program |
| 1398 | then computes the following distortion measures: |
| 1399 | .sp 0.2i |
| 1400 | hd2 - the magnitude of the frequency component 2*f1 assuming that f2 |
| 1401 | is not present. |
| 1402 | hd3 - the magnitude of the frequency component 3*f1 assuming that f2 |
| 1403 | is not present. |
| 1404 | sim2 - the magnitude of the frequency component f1 + f2. |
| 1405 | dim2 - the magnitude of the frequency component f1 - f2. |
| 1406 | dim3 - the magnitude of the frequency component 2*f1 - f2. |
| 1407 | .pp |
| 1408 | Rload is the name of the output load resistor into which all distortion |
| 1409 | power products are to be computed. Inter is the interval at which the summary |
| 1410 | printout of the contributions of all nonlinear devices to the total distortion |
| 1411 | is to be printed. If omitted or set to zero, no summary printout will be made. |
| 1412 | Refpwr is the reference power level used in computing the distortion products. |
| 1413 | if omitted, a value of 1 mw (that is, dbm) is used. Skw2 is the ratio of f2 to |
| 1414 | f1. If omitted, a value of 0.9 is used (i.e., f2 = 0.9*f1). Spw2 is the |
| 1415 | amplitude of f2. If omitted, a value of 1.0 is assumed. |
| 1416 | .pp |
| 1417 | The distortion measures hd2, hd3, sim2, dim2, and dim3 may also be be |
| 1418 | printed and/or plotted (see the description of the .print and .plot cards). |
| 1419 | .sp 0.2i |
| 1420 | .sh 2 ".noise card" |
| 1421 | .sp 0.2i |
| 1422 | .b "General form:" |
| 1423 | .(l |
| 1424 | .noise outv insrc nums |
| 1425 | .)l |
| 1426 | .b "Examples:" |
| 1427 | .(l |
| 1428 | .noise v(5) vin 10 |
| 1429 | .)l |
| 1430 | .pp |
| 1431 | This card controls the noise analysis of the circuit. The noise analysis |
| 1432 | is performed in conjunction with the ac analysis (see .ac card). Outv is an |
| 1433 | output voltage which defines the summing point. Insrc is the name of the |
| 1434 | independent voltage or current source which is the noise input reference. Nums |
| 1435 | is the summary interval. Spice will compute the equivalent output noise at |
| 1436 | the specified output as well as the equivalent input noise at the specified |
| 1437 | input. In addition, the contributions of every noise generator in the circuit |
| 1438 | will be printed at every nums frequency points (the summary interval). If nums |
| 1439 | is zero, no summary printout will be made. |
| 1440 | .pp |
| 1441 | The output noise and the equivalent input noise may also be printed and/or |
| 1442 | plotted (see the description of the .print and .plot cards). |
| 1443 | .sp 0.2i |
| 1444 | .sh 2 ".tran card" |
| 1445 | .sp 0.2i |
| 1446 | .b "General form:" |
| 1447 | .(l |
| 1448 | .tran tstep tstop <tstart <tmax>> <uic> |
| 1449 | .)l |
| 1450 | .b "Examples:" |
| 1451 | .(l |
| 1452 | .tran 1ns 100ns |
| 1453 | .tran 1ns 1000ns 500ns |
| 1454 | .tran 10ns 1us uic |
| 1455 | .)l |
| 1456 | .pp |
| 1457 | Tstep is the printing or plotting increment for line-printer output. |
| 1458 | For use with the post-processor, tstep is the suggested computing increment. |
| 1459 | tstop is the final time, and tstart is |
| 1460 | the initial time. If tstart is omitted, it is assumed to be zero. The |
| 1461 | transient analysis always begins at time zero. In the interval <zero, tstart>, |
| 1462 | the circuit is analyzed (to reach a steady state), but no outputs are stored. |
| 1463 | In the interval <tstart, tstop>, the circuit is analyzed and outputs are |
| 1464 | stored. Tmax is the maximum stepsize that spice will use (for default, the |
| 1465 | program chooses either tstep or (tstop-tstart)/50.0, whichever is smaller. |
| 1466 | Tmax is useful when one wishes too guarantee a computing interval which is |
| 1467 | smaller than the printer increment, tstep. |
| 1468 | .pp |
| 1469 | Uic (use initial conditions) is an optional keyword which indicates that |
| 1470 | the user does not want spice to solve for the quiescent operating point before |
| 1471 | beginning the transient analysis. If this keyword is specified, spice uses the |
| 1472 | values specified using ic=... On the various elements as the initial transient |
| 1473 | condition and proceeds with the analysis. If the .ic card has been specified, |
| 1474 | then the node voltages on the .ic card are used compute the intitial conditions |
| 1475 | for the devices. Look at the description on the .ic card for its |
| 1476 | interpretation when 'uic' is not specified. |
| 1477 | .sp 0.2i |
| 1478 | .sh 2 ".four card" |
| 1479 | .sp 0.2i |
| 1480 | .b "General form:" |
| 1481 | .(l |
| 1482 | .four freq ov1 <ov2 ov3 ...> |
| 1483 | .)l |
| 1484 | .b "Examples:" |
| 1485 | .(l |
| 1486 | .four 100k v(5) |
| 1487 | .)l |
| 1488 | .pp |
| 1489 | This card controls whether spice performs a fourier analysis as a part of |
| 1490 | the transient analysis. Freq is the fundamental frequency, and ov1, ..., are |
| 1491 | the output variables for which the analysis is desired. The fourier analysis |
| 1492 | is performed over the interval <tstop-period, tstop>, where tstop is the final |
| 1493 | time specified for the transient analysis, and period is one period of the |
| 1494 | fundamental frequency. The dc component and the first nine components are |
| 1495 | determined. For maximum accuracy, tmax (see the .tran card) should be set to |
| 1496 | period/100.0 (or less for very high-q circuits). |
| 1497 | .sp 0.2i |
| 1498 | .sh 2 ".print cards" |
| 1499 | .sp 0.2i |
| 1500 | .b "General form:" |
| 1501 | .(l |
| 1502 | .print prtype ov1 <ov2 ... Ov8> |
| 1503 | .)l |
| 1504 | .b "Examples:" |
| 1505 | .(l |
| 1506 | .print tran v(4) i(vin) |
| 1507 | .print ac vm(4,2) vr(7) vp(8,3) |
| 1508 | .print dc v(2) i(vsrc) v(23,17) |
| 1509 | .print noise inoise |
| 1510 | .print disto hd3 sim2(db) |
| 1511 | .)l |
| 1512 | .pp |
| 1513 | This card defines the contents of a tabular listing of one to eight output |
| 1514 | variables. Prtype is the type of the analysis (dc, ac, tran, noise, or |
| 1515 | distortion) for which the specified outputs are desired. The form for voltage o |
| 1516 | current output variables is as follows: |
| 1517 | .sp 0.2i |
| 1518 | .ip v(n1<,n2>) 10 |
| 1519 | specifies the voltage difference between nodes n1 |
| 1520 | and n2. If n2 (and the preceding comma) is omitted, |
| 1521 | ground (0) is assumed. For the ac analysis, five |
| 1522 | additional outputs can be accessed by replacing the |
| 1523 | letter v by: |
| 1524 | .sp 0.2i |
| 1525 | vr - real part |
| 1526 | vi - imaginary part |
| 1527 | vm - magnitude |
| 1528 | vp - phase |
| 1529 | vdb - 20*log10(magnitude) |
| 1530 | .sp 0.2i |
| 1531 | .ip i(vxxxxxxx) 10 |
| 1532 | specifies the current flowing in the independent |
| 1533 | voltage source named vxxxxxxx. Positive current |
| 1534 | flows from the positive node, through the source, to |
| 1535 | the negative node. For the ac analysis, the |
| 1536 | corresponding replacements for the letter i may be |
| 1537 | made in the same way as described for voltage outputs. |
| 1538 | .sp 0.2i |
| 1539 | .pp |
| 1540 | Output variables for the noise and distortion analyses have a different |
| 1541 | general form |
| 1542 | form from that of the other analyses. The is |
| 1543 | .(l |
| 1544 | ov<(x)> |
| 1545 | .)l |
| 1546 | where ov is any of onoise (output noise), inoise (equivalent input noise), |
| 1547 | hd2, hd3, sim2, dim2, or dim3 (see description of distortion analysis), and x |
| 1548 | may be any of: |
| 1549 | .(l |
| 1550 | r - real part |
| 1551 | i - imaginary part |
| 1552 | m - magnitude (default if nothing specified) |
| 1553 | p - phase |
| 1554 | db - 20*log10(magnitude) |
| 1555 | .)l |
| 1556 | thus, sim2 (or sim2(m)) describes the magnitude of the sim2 distortion measure, |
| 1557 | while hd2(r) describes the real part of the hd2 distortion measure. |
| 1558 | .pp |
| 1559 | There is no limit on the number of .print cards for each type of |
| 1560 | analysis. |
| 1561 | .sp 0.2i |
| 1562 | .sh 2 ".plot cards" |
| 1563 | .sp 0.2i |
| 1564 | .b "General form:" |
| 1565 | .(l |
| 1566 | .plot pltype ov1 <(plo1,phi1)> <ov2 <(plo2,phi2)> ... Ov8> |
| 1567 | .)l |
| 1568 | .b "Examples:" |
| 1569 | .(l |
| 1570 | .plot dc v(4) v(5) v(1) |
| 1571 | .plot tran v(17,5) (2,5) i(vin) v(17) (1,9) |
| 1572 | .plot ac vm(5) vm(31,24) vdb(5) vp(5) |
| 1573 | .plot disto hd2 hd3(r) sim2 |
| 1574 | .plot tran v(5,3) v(4) (0,5) v(7) (0,10) |
| 1575 | .)l |
| 1576 | .pp |
| 1577 | This card defines the contents of one plot of from one to eight output |
| 1578 | variables. Pltype is the type of analysis (dc, ac, tran, noise, or distortion) |
| 1579 | for which the specified outputs are desired. The syntax for the ovi is |
| 1580 | identical to that for the .print card, described above. |
| 1581 | .pp |
| 1582 | The optional plot limits (plo,phi) may be specified after any of the |
| 1583 | output variables. All output variables to the left of a pair of plot limits |
| 1584 | (plo,phi) will be plotted using the same lower and upper plot bounds. If plot |
| 1585 | limits are not specified, spice will automatically determine the minimum and |
| 1586 | maximum values of all output variables being plotted and scale the plot to fit. |
| 1587 | More than one scale will be used if the output variable values warrant (i.e., |
| 1588 | mixing output variables with values which are orders-of-magnitude different |
| 1589 | still gives readable plots). |
| 1590 | .pp |
| 1591 | The overlap of two or more traces on any plot is indicated by the letter |
| 1592 | x. |
| 1593 | .pp |
| 1594 | When more than one output variable appears on the same plot, the |
| 1595 | first variable specified will be printed as well as plotted. If a printout |
| 1596 | of all variables is desired, then a companion .print card should be included. |
| 1597 | .pp |
| 1598 | There is no limit on the number of .plot cards specified for each |
| 1599 | type of analysis. |
| 1600 | .bp |
| 1601 | .sh 1 "APPENDIX A: EXAMPLE DATA DECKS" |
| 1602 | .sp 0.2i |
| 1603 | .sh 2 "circuit 1" |
| 1604 | .pp |
| 1605 | The following deck determines the dc operating point and small-signal |
| 1606 | transfer function of a simple differential pair. In addition, the ac |
| 1607 | small-signal response is computed over the frequency range 1hz to 100meghz. |
| 1608 | .(l |
| 1609 | Simple differential pair |
| 1610 | Vcc 7 0 12 |
| 1611 | Vee 8 0 -12 |
| 1612 | Vin 1 0 ac 1 |
| 1613 | Rs1 1 2 1k |
| 1614 | Rs2 6 0 1k |
| 1615 | Q1 3 2 4 mod1 |
| 1616 | Q2 5 6 4 mod1 |
| 1617 | Rc1 7 3 10k |
| 1618 | Rc2 7 5 10k |
| 1619 | Re 4 8 10k |
| 1620 | .model mod1 npn bf=50 vbf=50 js=1.e-12 rb=100 cjc .5pf tf .6ns |
| 1621 | .tf v(5) vin |
| 1622 | .ac dec 10 1 100meg |
| 1623 | .plot ac vm(5) vp(5) |
| 1624 | .print ac vm(5) vp(5) |
| 1625 | .end |
| 1626 | .)l |
| 1627 | .sp 0.2i |
| 1628 | .sh 2 "circuit 2" |
| 1629 | .sp 0.2i |
| 1630 | The following deck computes the output characteristics of a mosfet |
| 1631 | device over the range 0-10v for vds and 0-5v for vgs. |
| 1632 | .sp 0.2i |
| 1633 | .(l |
| 1634 | Mos output characteristics |
| 1635 | .option nonode nopage |
| 1636 | Vds 3 0 |
| 1637 | Vgs 2 0 |
| 1638 | M1 1 2 0 0 mod1 l=4u w=6u ad=10p as=10p |
| 1639 | .model mod1 nmos vto=-2 nsub=1.0e15 uo=550 |
| 1640 | * vids measures id, we could have used vds, but id would be negative |
| 1641 | Vids 3 1 |
| 1642 | .dc vds 0 10 .5 vgs 0 5 1 |
| 1643 | .print dc i(vids) v(2) |
| 1644 | .plot dc i(vids) |
| 1645 | .end |
| 1646 | .)l |
| 1647 | .sp 0.2i |
| 1648 | .sh 2 "circuit 3" |
| 1649 | .sp 0.2i |
| 1650 | .pp |
| 1651 | The following deck determines the dc transfer curve and the transient |
| 1652 | pulse response of a simple rtl inverter. The input is a pulse from 0 to 5 |
| 1653 | volts with delay, rise, and fall times of 2ns and a pulse width of 30ns. The |
| 1654 | transient interval is 0 to 100ns, with printing to be done every nanosecond. |
| 1655 | .sp 0.2i |
| 1656 | .(l |
| 1657 | Simple rtl inverter |
| 1658 | Vcc 4 0 5 |
| 1659 | Vin 1 0 pulse 0 5 2ns 2ns 2ns 30ns |
| 1660 | Rb 1 2 10k |
| 1661 | Q1 3 2 0 q1 |
| 1662 | Rc 3 4 1k |
| 1663 | .plot dc v(3) |
| 1664 | .plot tran v(3) (0,5) |
| 1665 | .print tran v(3) |
| 1666 | .model q1 npn bf 20 rb 100 tf .1ns cjc 2pf |
| 1667 | .dc vin 0 5 0.1 |
| 1668 | .tran 1ns 100ns |
| 1669 | .end |
| 1670 | .)l |
| 1671 | .sp 0.2i |
| 1672 | .sh 2 "circuit 4" |
| 1673 | .pp |
| 1674 | The following deck simulates a four-bit binary adder, using several sub- |
| 1675 | circuits to describe various pieces of the overall circuit. |
| 1676 | .sp 0.2i |
| 1677 | .(l |
| 1678 | Adder - 4 bit all-nand-gate binary adder |
| 1679 | .sp 0.2i |
| 1680 | *** subcircuit definitions |
| 1681 | .sp 0.2i |
| 1682 | .subckt nand 1 2 3 4 |
| 1683 | * nodes: input(2), output, vcc |
| 1684 | Q1 9 5 1 qmod |
| 1685 | D1clamp 0 1 dmod |
| 1686 | Q2 9 5 2 qmod |
| 1687 | D2clamp 0 2 dmod |
| 1688 | Rb 4 5 4k |
| 1689 | R1 4 6 1.6k |
| 1690 | Q3 6 9 8 qmod |
| 1691 | R2 8 0 1k |
| 1692 | Rc 4 7 130 |
| 1693 | Q4 7 6 10 qmod |
| 1694 | Dvbedrop 10 3 dmod |
| 1695 | Q5 3 8 0 qmod |
| 1696 | .ends nand |
| 1697 | .subckt onebit 1 2 3 4 5 6 |
| 1698 | * nodes: input(2), carry-in, output, carry-out, vcc |
| 1699 | X1 1 2 7 6 nand |
| 1700 | X2 1 7 8 6 nand |
| 1701 | X3 2 7 9 6 nand |
| 1702 | X4 8 9 10 6 nand |
| 1703 | X5 3 10 11 6 nand |
| 1704 | X6 3 11 12 6 nand |
| 1705 | X7 10 11 13 6 nand |
| 1706 | X8 12 13 4 6 nand |
| 1707 | X9 11 7 5 6 nand |
| 1708 | .ends onebit |
| 1709 | .subckt twobit 1 2 3 4 5 6 7 8 9 |
| 1710 | * nodes: input - bit0(2) / bit1(2), output - bit0 / bit1, |
| 1711 | * carry-in, carry-out, vcc |
| 1712 | X1 1 2 7 5 10 9 onebit |
| 1713 | X2 3 4 10 6 8 9 onebit |
| 1714 | .ends twobit |
| 1715 | .sp 0.2i |
| 1716 | .subckt fourbit 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 |
| 1717 | * nodes: input - bit0(2) / bit1(2) / bit2(2) / bit3(2), |
| 1718 | * output - bit0 / bit1 / bit2 / bit3, carry-in, carry-out, vcc |
| 1719 | X1 1 2 3 4 9 10 13 16 15 twobit |
| 1720 | X2 5 6 7 8 11 12 16 14 15 twobit |
| 1721 | .ends fourbit |
| 1722 | .sp 0.2i |
| 1723 | *** define nominal circuit |
| 1724 | .sp 0.2i |
| 1725 | .model dmod d |
| 1726 | .model qmod npn(bf=75 rb=100 cje=1pf cjc=3pf) |
| 1727 | Vcc 99 0 dc 5v |
| 1728 | Vin1a 1 0 pulse(0 3 0 10ns 10ns 10ns 50ns) |
| 1729 | .sp 0.2i |
| 1730 | .sp 0.2i |
| 1731 | Vin1b 2 0 pulse(0 3 0 10ns 10ns 20ns 100ns) |
| 1732 | Vin2a 3 0 pulse(0 3 0 10ns 10ns 40ns 200ns) |
| 1733 | Vin2b 4 0 pulse(0 3 0 10ns 10ns 80ns 400ns) |
| 1734 | Vin3a 5 0 pulse(0 3 0 10ns 10ns 160ns 800ns) |
| 1735 | Vin3b 6 0 pulse(0 3 0 10ns 10ns 320ns 1600ns) |
| 1736 | Vin4a 7 0 pulse(0 3 0 10ns 10ns 640ns 3200ns) |
| 1737 | Vin4b 8 0 pulse(0 3 0 10ns 10ns 1280ns 6400ns) |
| 1738 | X1 1 2 3 4 5 6 7 8 9 10 11 12 0 13 99 fourbit |
| 1739 | Rbit0 9 0 1k |
| 1740 | Rbit1 10 0 1k |
| 1741 | Rbit2 11 0 1k |
| 1742 | Rbit3 12 0 1k |
| 1743 | Rcout 13 0 1k |
| 1744 | .plot tran v(1) v(2) v(3) v(4) v(5) v(6) v(7) v(8) |
| 1745 | .plot tran v(9) v(10) v(11) v(12) v(13) |
| 1746 | .print tran v(1) v(2) v(3) v(4) v(5) v(6) v(7) v(8) |
| 1747 | .print tran v(9) v(10) v(11) v(12) v(13) |
| 1748 | .sp 0.2i |
| 1749 | .tran 1ns 6400ns |
| 1750 | *** (for those with money (and memory) to burn) |
| 1751 | .sp 0.2i |
| 1752 | .opt acct list node limpts=6401 |
| 1753 | .end |
| 1754 | .)l |
| 1755 | .sp 0.2i |
| 1756 | .sh 2 "circuit 5" |
| 1757 | .pp |
| 1758 | The following deck simulates a transmission-line inverter. Two |
| 1759 | transmission-line elements are required since two propagation modes are excited. |
| 1760 | In the case of a coaxial line, the first line (t1) models the inner conductor wi |
| 1761 | respect to the shield, and the second line (t2) models the shield with respect |
| 1762 | to the outside world. |
| 1763 | .sp 0.2i |
| 1764 | .(l |
| 1765 | Transmission-line inverter |
| 1766 | V1 1 0 pulse(0 1 0 0.1n) |
| 1767 | R1 1 2 50 |
| 1768 | X1 2 0 0 4 tline |
| 1769 | R2 4 0 50 |
| 1770 | .subckt tline 1 2 3 4 |
| 1771 | T1 1 2 3 4 z0=50 td=1.5ns |
| 1772 | T2 2 0 4 0 z0=100 td=1ns |
| 1773 | .ends tline |
| 1774 | .tran 0.1ns 20ns |
| 1775 | .plot tran v(2) v(4) |
| 1776 | .end |
| 1777 | .)l |
| 1778 | .bp |
| 1779 | .sh 1 "APPENDIX B: NONLINEAR DEPENDENT SOURCES" |
| 1780 | .pp |
| 1781 | Spice allows circuits to contain dependent sources characterized by any of |
| 1782 | the four equations |
| 1783 | .sp 0.2i |
| 1784 | i=f(v) v=f(v) i=f(i) v=f(i) |
| 1785 | .sp 0.2i |
| 1786 | where the functions must be polynomials, and the arguments may be |
| 1787 | multidimensional. The polynomial functions are specified by a set of coefficien |
| 1788 | p0, p1, ..., pn. Both the number of dimensions and the number of coefficients |
| 1789 | are arbitrary. The meaning of the coefficients depends upon the dimension of |
| 1790 | the polynomial, as shown in the following examples: |
| 1791 | .pp |
| 1792 | Suppose that the function is one-dimensional (that is, a function of one |
| 1793 | argument). Then the function value fv is determined by the following |
| 1794 | expression in fa (the function argument): |
| 1795 | .sp 0.2i |
| 1796 | fv = p0 + (p1*fa) + (p2*fa**2) + (p3*fa**3) + (p4*fa**4) |
| 1797 | .sp 0.2i |
| 1798 | + (p5*fa**5) + ... |
| 1799 | .pp |
| 1800 | Suppose now that the function is two-dimensional, with arguments fa and |
| 1801 | fb. Then the function value fv is determined by the following expression: |
| 1802 | .sp 0.2i |
| 1803 | fv = p0 + (p1*fa) + (p2*fb) + (p3*fa**2) + (p4*fa*fb) + (p5*fb**2) |
| 1804 | .sp 0.2i |
| 1805 | + (p6*fa**3) + (p7*fa**2*fb) + (p8*fa*fb**2) + (p9*fb**3) + ... |
| 1806 | .pp |
| 1807 | Consider now the case of a three-dimensional polynomial function with |
| 1808 | arguments fa, fb, and fc. Then the function value fv is determined by the |
| 1809 | following expression: |
| 1810 | .sp 0.2i |
| 1811 | fv = p0 + (p1*fa) + (p2*fb) + (p3*fc) + (p4*fa**2) + (p5*fa*fb) |
| 1812 | .sp 0.2i |
| 1813 | + (p6*fa*fc) + (p7*fb**2) + (p8*fb*fc) + (p9*fc**2) + (p10*fa**3) |
| 1814 | .sp 0.2i |
| 1815 | + (p11*fa**2*fb) + (p12*fa**2*fc) + (p13*fa*fb**2) |
| 1816 | .sp 0.2i |
| 1817 | + (p14*fa*fb*fc) |
| 1818 | .sp 0.2i |
| 1819 | + (p15*fa*fc**2) + (p16*fb**3) + (p17*fb**2*fc) + (p18*fb*fc**2) |
| 1820 | .sp 0.2i |
| 1821 | + (p19*fc**3) + (p20*fa**4) + ... |
| 1822 | .pp |
| 1823 | Note: if the polynomial is one-dimensional and exactly one coefficient is |
| 1824 | specified, then spice assumes it to be p1 (and p0 = 0.0), in order to |
| 1825 | facilitate the input of linear controlled sources. |
| 1826 | .pp |
| 1827 | For all four of the dependent sources described below, the initial |
| 1828 | condition parameter is described as optional. If not specified, spice assumes 0 |
| 1829 | the initial condition for dependent sources is an initial 'guess' for the value |
| 1830 | of the controlling variable. The program uses this initial condition to obtain |
| 1831 | the dc operating point of the circuit. After convergence has been obtained, |
| 1832 | the program continues iterating to obtain the exact value for the controlling |
| 1833 | variable. Hence, to reduce the computational effort for the dc operating |
| 1834 | point (or if the polynomial specifies a strong nonlinearity), a value fairly |
| 1835 | close to the actual controlling variable should be specified for the initial |
| 1836 | condition. |
| 1837 | .sh 2 "voltage-controlled current sources" |
| 1838 | .sp 0.2i |
| 1839 | .b "General form:" |
| 1840 | .(l |
| 1841 | gxxxxxxx n+ n- <poly(nd)> nc1+ nc1- ... P0 <p1 ...> <ic=...> |
| 1842 | .)l |
| 1843 | .sp 0.2i |
| 1844 | Examples: g1 1 0 5 3 0 0.1mmho |
| 1845 | gr 17 3 17 3 0 1m 1.5m ic=2v |
| 1846 | gmlt 23 17 poly(2) 3 5 1 2 0 1m 17m 3.5u ic=2.5, 1.3 |
| 1847 | .pp |
| 1848 | N+ and n- are the positive and negative nodes, respectively. Current flow |
| 1849 | is from the positive node, through the source, to the negative node. Poly(nd) |
| 1850 | only has to be specified if the source is multi-dimensional (one-dimensional is |
| 1851 | the default). If specified, nd is the number of dimensions, which must be |
| 1852 | positive. Nc1+, nc1-, ... Are the positive and negative controlling nodes, |
| 1853 | respectively. One pair of nodes must be specified for each dimension. P0, p1, |
| 1854 | p2, ..., pn are the polynomial coefficients. The (optional) initial condition |
| 1855 | is the initial guess at the value(s) of the controlling voltage(s). If not |
| 1856 | specified, 0.0 is assumed. The polynomial specifies the source current as a |
| 1857 | function of the controlling voltage(s). The second example above describes a |
| 1858 | current source with value |
| 1859 | .sp 0.2i |
| 1860 | i = 1e-3*v(17,3) + 1.5e-3*v(17,3)**2 |
| 1861 | .sp 0.2i |
| 1862 | note that since the source nodes are the same as the controlling nodes, this |
| 1863 | source actually models a nonlinear resistor. |
| 1864 | .sp 0.2i |
| 1865 | .sp 0.2i |
| 1866 | .sp 0.2i |
| 1867 | .sp 0.2i |
| 1868 | .sp 0.2i |
| 1869 | .sh 2 "voltage-controlled voltage sources" |
| 1870 | .sp 0.2i |
| 1871 | .b "General form:" |
| 1872 | .(l |
| 1873 | exxxxxxx n+ n- <poly(nd)> nc1+ nc1- ... P0 <p1 ...> <ic=...> |
| 1874 | .)l |
| 1875 | .sp 0.2i |
| 1876 | Examples: e1 3 4 21 17 10.5 2.1 1.75 |
| 1877 | ex 17 0 poly(3) 13 0 15 0 17 0 0 1 1 1 ic=1.5,2.0,17.35 |
| 1878 | .pp |
| 1879 | N+ and n- are the positive and negative nodes, respectively. Poly(nd) |
| 1880 | only has to be specified if the source is multi-dimensional (one-dimensional is |
| 1881 | the default). If specified, nd is the number of dimensions, which must be |
| 1882 | positive. Nc1+, nc1-, ... Are the positive and negative controlling nodes, |
| 1883 | respectively. One pair of nodes must be specified for each dimension. P0, p1, |
| 1884 | p2, ..., pn are the polynomial coefficients. The (optional) initial condition |
| 1885 | is the initial guess at the value(s) of the controlling voltage(s). If not |
| 1886 | specified, 0.0 is assumed. The polynomial specifies the source voltage as a |
| 1887 | function of the controlling voltage(s). The second example above describes a |
| 1888 | voltage source with value |
| 1889 | .sp 0.2i |
| 1890 | v = v(13,0) + v(15,0) + v(17,0) |
| 1891 | .sp 0.2i |
| 1892 | (in other words, an ideal voltage summer). |
| 1893 | .sh 2 "current-controlled current sources" |
| 1894 | .sp 0.2i |
| 1895 | .b "General form:" |
| 1896 | .(l |
| 1897 | fxxxxxxx n+ n- <poly(nd)> vn1 <vn2 ...> p0 <p1 ...> <ic=...> |
| 1898 | .)l |
| 1899 | .sp 0.2i |
| 1900 | Examples: f1 12 10 vcc 1ma 1.3m |
| 1901 | fxfer 13 20 vsens 0 1 |
| 1902 | .pp |
| 1903 | N+ and n- are the positive and negative nodes, respectively. Current flow |
| 1904 | is from the positive node, through the source, to the negative node. Poly(nd) |
| 1905 | only has to be specified if the source is multi-dimensional (one-dimensional is |
| 1906 | the default). If specified, nd is the number of dimensions, which must be |
| 1907 | positive. Vn1, vn2, ... Are the names of voltage sources through which the |
| 1908 | controlling current flows; one name must be specified for each dimension. The |
| 1909 | direction of positive controlling current flow is from the positive node, |
| 1910 | through the source, to the negative node of each voltage source. P0, p1, |
| 1911 | p2, ..., pn are the polynomial coefficients. The (optional) initial condition |
| 1912 | is the initial guess at the value(s) of the controlling current(s) (in amps). |
| 1913 | If not specified, 0.0 is assumed. The polynomial specifies the source current |
| 1914 | as a function of the controlling current(s). The first example above describes |
| 1915 | a current source with value |
| 1916 | .sp 0.2i |
| 1917 | i = 1e-3 + 1.3e-3*i(vcc) |
| 1918 | .sp 0.2i |
| 1919 | .sp 0.2i |
| 1920 | .sp 0.2i |
| 1921 | .sp 0.2i |
| 1922 | .sp 0.2i |
| 1923 | .sh 2 "current-controlled voltage sources" |
| 1924 | .sp 0.2i |
| 1925 | .b "General form:" |
| 1926 | .(l |
| 1927 | hxxxxxxx n+ n- <poly(nd)> vn1 <vn2 ...> p0 <p1 ...> <ic=...> |
| 1928 | .)l |
| 1929 | .sp 0.2i |
| 1930 | Examples: hxy 13 20 poly(2) vin1 vin2 0 0 0 0 1 ic=0.5 1.3 |
| 1931 | hr 4 17 vx 0 0 1 |
| 1932 | .pp |
| 1933 | N+ and n- are the positive and negative nodes, respectively. Poly(nd) |
| 1934 | only has to be specified if the source is multi-dimensional (one-dimensional is |
| 1935 | the default). If specified, nd is the number of dimensions, which must be |
| 1936 | positive. Vn1, vn2, ... Are the names of voltage sources through which the |
| 1937 | controlling current flows; one name must be specified for each dimension. The |
| 1938 | direction of positive controlling current flow is from the positive node, |
| 1939 | through the source, to the negative node of each voltage source. P0, p1, |
| 1940 | p2, ..., pn are the polynomial coefficients. The (optional) initial condition |
| 1941 | is the initial guess at the value(s) of the controlling current(s) (in amps). |
| 1942 | If not specified, 0.0 is assumed. The polynomial specifies the source voltage |
| 1943 | as a function of the controlling current(s). The first example above describes |
| 1944 | a voltage source with value |
| 1945 | .sp 0.2i |
| 1946 | v = i(vin1)*i(vin2) |